Systems and method for orthogonal frequency divisional multiplexing

ABSTRACT

Systems and methods for orthogonal frequency division multiplexing are provided. In one embodiment, a multi-carrier modem comprises: a receiver configured to de-modulate symbols from at least one of a plurality of spectrally overlapping carrier signals to produce a receiver output; a transmitter configured to modulate symbols onto at least one of a plurality of the spectrally overlapping carrier signals; a processor coupled to the transmitter, wherein the processor outputs data for transmission by the transmitter, wherein the processor applies an inverse Fourier transform to the data transmitted by the transmitter; the processor coupled to the receiver, wherein the processor applies a Fourier transform to the receiver output; and a controller programmed to instruct the transmitter to transmit at least one symbol representing a request for bandwidth allocation on a first carrier; wherein the controller is further programmed to determine when a collision has occurred on the first carrier.

CROSS REFERENCE TO RELATED CASES

This application is a divisional of application Ser. No. 13/707,343filed Dec. 6, 2012, which is a divisional of application Ser. No.11/927,405 filed Oct. 29, 2007, entitled “SYSTEMS AND METHOD FORORTHOGONAL FREQUENCY DIVISIONAL MULTIPLEXING”, which is a continuationof application Ser. No. 11/420,851 filed on May 30, 2006, entitled“DYNAMIC BANDWIDTH ALLOCATION” (currently pending), which is adivisional of application Ser. No. 09/903,273 filed Jul. 11, 2001,entitled “DYNAMIC BANDWIDTH ALLOCATION” (now U.S. Pat. No. 7,069,577),which is a continuation of application Ser. No. 09/397,443, filed Sep.15, 1999, entitled “DYNAMIC BANDWIDTH ALLOCATION” (now U.S. Pat. No.6,279,158), which is a divisional of U.S. application Ser. No.08/673,002 filed Jun. 28, 1996 (now U.S. Pat. No. 6,334,219) which is acontinuation-in-part of U.S. application Ser. No. 08/650,408 filed May20, 1996 (abandoned), Ser. No. 08/457,295 filed Jun. 1, 1995(abandoned), Ser. No. 08/457,317 filed Jun. 1, 1995 (abandoned), Ser.No. 08/311,964 filed Sep. 26, 1994 (abandoned), and Ser. No. 08/384,659filed Feb. 6, 1995 (abandoned) whose applications are incorporatedherein by reference.

This application is related to U.S. application Ser. No. 08/455,340filed May 31, 1995 (abandoned), Ser. No. 08/455,059 filed May 31, 1995(abandoned), Ser. No. 08/457,294 filed Jun. 1, 1995 (abandoned), Ser.No. 08/457,110 filed Jun. 1, 1995 (abandoned), Ser. No. 08/456,871 filedJun. 1, 1995 (abandoned), Ser. No. 08/457,022 filed Jun. 1, 1995(abandoned), and Ser. No. 08/457,037 filed Jun. 1, 1995 (abandoned),whose applications are incorporated herein by reference.

BACKGROUND

Information services found in households and businesses today includetelevision (or video) services and telephone services. Anotherinformation service involves digital data transfer which is mostfrequently accomplished using a modem connected to a telephone service.All further references to telephony herein shall include both telephoneservices and digital data transfer service.

Characteristics of telephony and video signals are different andtherefore telephony and video networks are designed differently as well.For example, telephony information occupies a relatively narrow bandwhen compared to the bandwidth for video signals. In addition, telephonysignals are low frequency whereas NTSC standard video signals aretransmitted at carrier frequencies greater than 50 MHz. Accordingly,telephone transmission networks are relatively narrow band systems whichoperate at audio frequencies and which typically serve the customer bytwisted wire drops from a curb-side junction box. On the other hand,cable television services are broad band and incorporate variousfrequency carrier mixing methods to achieve signals compatible withconventional very high frequency television receivers. Cable televisionsystems or video services are typically provided by cable televisioncompanies through a shielded cable service connection to each individualhome or business.

One attempt to combine telephony and video services into a singlenetwork is described in U.S. Pat. No. 4,977,593 to Balance entitled“Optical Communications Network.” Balance describes a passive opticalcommunications network with an optical source located in a centralstation. The optical source transmits time division multiplexed opticalsignals along an optical fiber and which signals are later split by aseries of splitters between several individual fibers servicingoutstations. The network allows for digital speech data to betransmitted from the outstations to the central station via the sameoptical path. In addition, Balance indicates that additional wavelengthscould be utilized to add services, such as cable television, via digitalmultiplex to the network.

A 1988 NCTA technical paper, entitled “Fiber Backbone: A Proposal For anEvolutionary Cable TV network Architecture,” by James A. Chiddix andDavid M. Pangrac, describes a hybrid optical fiber/coaxial cabletelevision (CATV) system architecture. The architecture builds uponexisting coaxial CATV networks. The architecture includes the use of adirect optical fiber path from a head end to a number of feed points inan already existing CATV distribution system.

U.S. Pat. No. 5,153,763 to Pidgeon, entitled “CATV Distribution NetworksUsing Light Wave Transmission Lines,” describes a CATV network fordistribution of broad band, multichannel CATV signals from a head end toa plurality of subscribers. Electrical to optical transmitters at thehead end and optical to electrical receivers at a fiber node launch andreceive optical signals corresponding to broad band CATV electricalsignals. Distribution from the fiber node is obtained by transmittingelectrical signals along coaxial cable transmission lines. The systemreduces distortion of the transmitted broad band CATV signals by blockconversion of all or part of the broad band of CATV signals to afrequency range which is less than an octave. Related U.S. Pat. No.5,262,883 to Pidgeon, entitled “CATV Distribution Networks Using LightWave Transmission Lines,” further describes the distortion reducingsystem.

Although the above-mentioned networks describe various concepts fortransmitting broad band video signals over various architectures, whichmay include hybrid optical fiber/coax architectures, none of thesereferences describe a cost effective, flexible, communications systemfor telephony communications. Several problems are inherent in such acommunication system.

One such problem is the need to optimize the bandwidth used fortransporting data so that the bandwidth used does not exceed theallotted bandwidth. Bandwidth requirements are particularly critical inmulti-point to point communication where multiple transmitters at remoteunits must be accommodated such that allotted bandwidth is not exceeded.

A second problem involves power consumption of the system. Thecommunication system should minimize the power used at the remote unitsfor the transport of data, as the equipment utilized at the remote unitsfor transmission and reception may be supplied by power distributed overthe transmission medium of the system.

Another problem arises from a fault in the system preventingcommunication between a head end and multiple remote units of amulti-point to point system. For example, a cut transmission line from ahead end to many remote units may leave many users without service.After the fault is corrected, it is important to bring as many remoteunits back into service as quickly as possible.

Data integrity must also be addressed. Both internal and externalinterference can degrade the communication. Internal interference existsbetween data signals being transported over the system. That is,transported data signals over a common communication link may experienceinterference there between, decreasing the integrity of the data.Ingress from external sources can also effect the integrity of datatransmissions. A telephony communication network is susceptible to“noise” generated by external sources, such as HAM radio. Because suchnoise can be intermittent and vary in intensity, a method oftransporting data over the system should correct or avoid the presenceof such ingress.

These problems, and others as will become apparent from the descriptionto follow, present a need for an enhanced communication system.Moreover, once the enhanced system is described, a number of practicalproblems in its physical realization are presented and overcome.

Another embodiment provides a method and apparatus for a fast Fouriertransform (FFT). This invention relates to the field of electroniccommunication systems, and more specifically to an improved method andapparatus for providing an FFT.

There are many advanced digital signal-processing applications requiringanalysis of large quantities of data in short time periods, especiallywhere there is interest in providing “real time” results. Suchapplications include signal processing in modems which use OFDM(orthogonal frequency division multiplexing). In order to be useful inthese and other applications, Discrete Fourier Transform (DFT) or FFTsignal processors must accommodate large numbers of transforms, oramounts of data, in very short processing times, often called high datathroughput.

In addition to the speed and data-throughput requirements, powerconsumption is a major concern for many applications. In somesignal-processing applications, power is supplied by portable generationor storage equipment, such as batteries, where the ultimate poweravailable is limited by many environment. In such applications,processor power consumption must be as low as possible. One usefulmeasure of utility or merit for FFT processors is the energy dissipationper transform point. Ultimately, one key problem with any FFT processoris the amount of power consumed per transform. Generally,high-performance, efficient FFT processors exhibit energy dissipationsper transform in the range of 100 to 1000 times log 2N nanojoules, whereN is the number of points in a given transform. As a consequence,reasonably large transforms required to process large arrays of data,result in large power consumption.

Machine-implemented computation of an FFT is often simplified bycascading together a series of simple multiply-and-add stages. When arecursive process is used, data circulates through a single stage andthe computational structure of the stage is made variable for eachcirculation. Each circulation through the stage is referred to as a“pass.”

A plurality of computational elements, each known as a radix-rbutterfly, may be assembled to define a single stage for carrying out aparticular pass. A radix-r butterfly receives r input signals andproduces a corresponding number of r output signals, where each outputsignal is the weighted sum of the r input signals. The radix number, r,in essence, defines the number of input components which contribute toeach output component.

By way of example, a radix-2 butterfly receives two input signals andproduces two output signals. Each output signal is the weighted sum ofthe two input signals. A radix-4 butterfly receives four input signalsand produces four corresponding output signals. Each output signal ofthe radix-4 butterfly constitutes a weighted sum of the four inputsignals.

Completion of an N-point FFT requires that the product of the butterflyradix values, taken over the total number of stages or passes, equalsthe total point count, N. Thus, a 64-point FFT can be performed by oneradix-64 butterfly, or three cascaded stages where each stage hassixteen radix-4 butterflies (the product of the radix values for stage-1and stage-2 and stage-3 is 4×4×4=64), or six cascaded stages where eachof the six stages comprises 32 radix-2 butterflies (the product of theradix values for stage-1 through stage-6 is 2×2×2×2×2×2=64).

A multi-stage or multi-pass FFT process can be correctly carried outunder conditions where the number of butterfly elements changes from onepass (or stage) to the next and the radix value, r, of the butterflyelements also changes from one pass (or stage) to the next. A paper byGordon DeMuth, “ALGORITHMS FOR DEFINING MIXED RADIX FFT FLOW GRAPHS,”IEEE Transactions on Acoustics, Speech, and Signal Processing, Vol. 37,No. 9, September 1989, Pages 1349-1358, describes a generalized methodfor performing an FFT with a mixed-radix system. A mixed-radix system isone where the radix value, r, in one stage or pass is different fromthat of at least one other stage or pass.

An advantage of a mixed-radix computing system is that it can be “tuned”to optimize the signal-to-noise ratio of the transform (or morecorrectly speaking, to minimize the accumulated round-off error of thetotal transform) for each particular set of circumstances. By way ofexample, it is advantageous in one environment to perform a 512-pointFFT using the mixed-radix sequence: 4, 4, 4, 4, 2. In a differentenvironment, it may be more advantageous to use the mixed-radixsequence: 4, 2, 4, 4, 4. Round-off error varies within a machine offinite precision as a function of radix value and the peak signalmagnitudes that develop in each stage or pass.

In addition, it may be advantageous to scale intermediate resultsbetween each stage or pass, in order to minimize round-off errors andthe problem of overflow. Further, it may be advantageous to vary theamount of scaling performed between each pass, for example, either toscale by ¼ between each radix-4 stage or to scale by ½ for some stagesand ⅛ for other stages.

Heretofore, FFT processors generally fetched data values from theirworking storage in a serial manner, thus limiting the speed which couldbe obtained. Further, current FFT processors generally were limited inspeed by loading the working storage with input values, then processingthe data in the working storage, then unloading the result values.

There are many advanced digital signal-processing applications requiringanalysis of large quantities of data in short time periods, especiallywhere there is interest in providing “real time” results. Suchapplications include signal processing in modems which use OFDM(orthogonal frequency division multiplexing).

One need in the art is for an accurate analog-to-digital conversion(ADC) at moderate frequencies having limited bandwidth. One technologyknown in the art is the “Sigma-Delta” ADC which provides very goodresolution (high number of bits in the digital result), but only forsignals whose converted signal bandwidth is low.

Another need is for an ADC which provides bandwidth-limited digital Iand Q signals (representing amplitude and quadrature) for a 200 kHzbandwidth received analog modem signal, wherein the digital result hasvery high resolution and accuracy.

What is needed is a method and apparatus which addresses the aboveproblems in the art.

SUMMARY

The present invention describes a multi-point to point communicationsystem including multi carrier telephony transport. The multi-point topoint communication system includes a hybrid fiber/coax distributionnetwork. A head end terminal provides for downstream transmission ofdownstream control data and downstream telephony information in a firstfrequency bandwidth over the hybrid fiber/coax distribution network andreception of upstream telephony information and upstream control data ina second frequency bandwidth over the hybrid fiber/coax distributionnetwork. The head end terminal includes a head end multicarrier modemfor modulating at least downstream telephony information on a pluralityof orthogonal carriers in the first frequency bandwidth and demodulatingat least upstream telephony information modulated on a plurality oforthogonal carriers in the second frequency bandwidth. The head endterminal further includes a head end controller operatively connected tothe head end multicarrier modem for controlling transmission of thedownstream telephony information and downstream control data and forcontrolling receipt of the upstream control data and upstream telephonyinformation. The system further includes at least one service unit, eachservice unit associated with at least one remote unit and operativelyconnected to the hybrid fiber/coax distribution network for upstreamtransmission of upstream telephony information and upstream control datain the second frequency bandwidth and for receipt of the downstreamcontrol data and downstream telephony information in the first frequencybandwidth. Each service unit includes a service unit multicarrier modemfor modulating at least the upstream telephony information on at leastone carrier orthogonal at the head end to at least one other carrier inthe second frequency bandwidth and for demodulating at least thedownstream telephony information modulated on at least a band of aplurality of orthogonal carriers in the first frequency bandwidth. Eachservice unit also includes a service unit controller operativelyconnected to the service unit multicarrier modem for controlling themodulation of and demodulation performed by the service unitmulticarrier modem.

Another embodiment prevents untoward spectral effects in themulticarrier signal from variations in channel activity and from highlyrepetitive data patterns in the payload channels. Data is the payloadchannels can be scrambled with pseudorandom sequences, and differentsequences can be applied to different channels in order to produce amore balanced multicarrier spectrum.

In another embodiment, the plurality of orthogonal carriers in the firstfrequency bandwidth include at least one control channel fortransmission of downstream control data and a plurality of telephonyinformation channels for transmission of downstream telephonyinformation. Further, the plurality of orthogonal carriers in the secondfrequency bandwidth include at least one control channel fortransmission of upstream control data and a plurality of telephonyinformation channels for transmission of upstream telephony information.

In other embodiments, a plurality of control channels are interspersedamong the telephony information channels in the first frequencybandwidth and a plurality of control channels are interspersed among thetelephony channels of the second frequency bandwidth. The telephonychannels may be divided into subbands each having multiple data orpayload channels and a control channel; this allows the remote modems tobe realized as less expensive and/or better performing narrow-bandmodems. The clock signals for generating the carriers and the symbolsrepresenting the transmitted data may be locked to each other orgenerated from the same source, to reduce intersymbol interferencesignificantly.

Another technique for reducing intersymbol interference is thetransmission of each symbol with more than 360° of phase in one cycle ofits carrier, in order to allow some leeway in tracking the phase of achannel carrier in a receiving modem. Some applications demand more ordifferent error detection and correction capability than others. Anembodiment is shown which handles both unencoded parity-typedetection/correction and more multiple types of more powerful methods,such as Reed-Solomon encoding, in a transparent, real-time fashion, bypacking the data words differently for each case. Moreover, theprocessor loading involved in these error-correction techniques can bespread out in time, so that not all channels need to be handled at thesame time. This is accomplished by staggering the beginning times ofdifferent data messages.

In another embodiment, the at least one service unit includes a servicemodem for upstream transmission of upstream telephony information andupstream control data within a channel band of the second frequencybandwidth corresponding to one of the channel bands of the firstfrequency bandwidth in which the service modem receives downstreamtelephony information and downstream control information. Alternatively,the at least one service unit includes a multi-service modem forupstream transmission of upstream telephony information and upstreamcontrol data within a plurality of channel bands of the second frequencybandwidth corresponding to a plurality of the channel bands of the firstfrequency bandwidth in which the multi-service modem receives downstreamtelephony information and downstream control information.

In still another embodiment, the plurality of control channels of thefirst frequency bandwidth and the plurality of control channels of thesecond frequency bandwidth each include at least one synchronizationchannel.

In other embodiments, different modulation techniques are utilized fordifferent carriers. For example, different modulation techniques areutilized for different telephony channels. As another example, theaforementioned IOC channels may be modulated as differential binaryphase-shift keyed (BPSK) signals, while the payload data channels aremodulated as 5-bit quadrature amplitude modulated (QAM32) signals, inorder to enhance the use of IOC channels for subband tracking, and forother purposes. The constellation defining the modulated signals can beconstructed to achieve a minimal number of bit errors for small errorsin amplitude or phase of the received signal; broadly, the constellationpoints are mapped to bit combinations in a scheme analogous to a Graycode.

A communication system which addresses the problems inherent in thesystem, in particular, ingress problems is also described. Thecommunication system includes a distribution network between a head endterminal and at least one remote unit. The head end terminal receivesupstream telephony information and upstream control data in a frequencybandwidth over the distribution network. The head end terminal includesa head end multicarrier demodulator for demodulating at least upstreamtelephony information modulated on a plurality of orthogonal carriers inthe frequency bandwidth. The demodulator includes at least one polyphasefilter for filtering the at least upstream telephony informationmodulated on the plurality of orthogonal carriers to provide ingressprotection for the modulated orthogonal carriers. The head end terminalalso includes a head end controller operatively connected to the headend multicarrier demodulator for controlling receipt of the upstreamcontrol data and upstream telephony information. The system furtherincludes at least one service unit modulator, each service unitmodulator associated with at least one remote unit and operativelyconnected to the distribution network for modulating at least upstreamtelephony information on at least one carrier orthogonal at the head endterminal to at least one other carrier in the frequency bandwidth. Thesystem also includes a service unit controller operatively connected tothe service unit multicarrier modulator for controlling the modulationperformed by the service unit multicarrier modulator.

In another embodiment, the plurality of orthogonal carriers in thefrequency bandwidth include a plurality of telephony informationchannels for transmission of upstream telephony information aftermodulation of telephony information thereon and at least one controlchannel associated with the plurality of telephony channels fortransmission of upstream control data thereon. Here also, the IOC may beplaced in the midpoint of the subbands.

In another embodiment, the at least one polyphase filter includes afirst and second polyphase filter. The first polyphase filter filters afirst plurality of channel sets and passes a first plurality of at leasttelephony channels within each channel set of the first plurality ofchannel sets. The second polyphase filter filters a second plurality ofchannel sets and passes a second plurality of at least telephonychannels within each channel set of the second plurality of channelsets. The first and second polyphase filter are offset from one anothersuch that all at least telephony channels of the first and secondplurality of channel sets are passed. In another embodiment, thepolyphase filters include at least two overlapping polyphase filters.

In another alternate embodiment, the demodulator includes a tunablenotch filter for filtering the at least upstream telephony informationmodulated on a plurality of orthogonal carriers to prevent passage ofcorrupted modulated orthogonal carriers.

In addition, a method of polyphase filtering in a communication systemis also described. The method includes receiving a plurality oforthogonal carriers having modulated telephony information thereon. Theplurality of orthogonal carriers includes a first and second pluralityof noncontiguous channel sets. The first plurality of noncontiguouschannel sets is filtered and a first plurality of channels of eachchannel set of the first plurality of noncontiguous channel sets ispassed. The second plurality of noncontiguous channel sets is filteredand a second plurality of channels of each channel set of the secondplurality of noncontiguous channel sets is also passed. The secondplurality of channels passed includes channels of the first plurality ofnoncontiguous channel sets not passed when filtering the first pluralityof noncontiguous channel sets.

A receiver apparatus is also described which receives a frequencybandwidth having a plurality of modulated orthogonal carriers. At leastone polyphase filter provides ingress protection for the frequencybandwidth by filtering a plurality of channel sets of the modulatedorthogonal carriers.

The use of channel monitoring to address some of the problems inherentin a multi-point to point communication system, in particular, withrespect to ingress, is also described. The monitoring method of thepresent invention monitors a telephony communication n-bit channelwherein one of the bits is a parity bit. The parity bit of the n-bitchannel is sampled and a probable bit error rate is derived from thesampling of the parity bit.

In one embodiment, the probable bit error rate over a time period iscompared to a predetermined bit error rate value representing a minimumbit error rate to determine if the n-bit channel is corrupted. Acorrupted channel can then either be reallocated or, in anotherembodiment, the transmission power of the channel can be increased toovercome the corruption.

In an alternate method embodiment, the method comprises the steps ofsampling the parity bit of the n-bit channel over a first time period,deriving a probable bit error rate from the sampling of the parity bitover the first time period, comparing the probable bit error rate overthe first time period to a pre-determined bit error rate value todetermine if the n-bit channel is corrupted, and accumulating a probablebit error rate over a plurality of successive time periods if the n-bitchannel is not corrupted.

In another alternate method embodiment, the method comprises the stepsof sampling the parity bit of the n-bit channel and deriving a probablebit error rate from the sampling of the parity bit over a first timeperiod. The probable bit error rate over the first time period iscompared to a first predetermined bit error rate value to determine ifthe n-bit channel is corrupted. A probable bit error rate from thesampling of the parity bit over a second time period is derived. Thesecond time period is longer than the first time period and runsconcurrently therewith. The probable bit error rate over the second timeperiod is compared to a second predetermined bit error rate value todetermine if the n-bit channel is corrupted.

In yet another alternate embodiment, a method for monitoring at leastone telephony communication channel includes equalizing a signal on thechannel and monitoring the equalization of the signal to produce aprobable bit error rate as a function of the equalization.

In still yet another alternate embodiment, a method for monitoring atleast one unallocated telephony communication channel includesperiodically monitoring the at least one unallocated telephonycommunication channel. Error data for the at least one unallocatedtelephony communication channel accumulated and the at least oneunallocated telephony communication channel is allocated based on theerror data.

A multi-point to point communication system utilizing a distributed loopmethod is also described. The communication system in accordance withthe present invention includes a distribution network and a head endterminal for downstream transmission of downstream control data anddownstream telephony information in a first frequency bandwidth over thedistribution network. The head end terminal receives upstream telephonyinformation and upstream control data in a second frequency bandwidthover the distribution network. The head end terminal further includes ahead end multi carrier modem for modulating at least downstreamtelephony information on a plurality of orthogonal carriers in the firstfrequency bandwidth and demodulating at least upstream telephonyinformation modulated on a plurality of orthogonal carriers in thesecond frequency bandwidth. A head end controller is operativelyconnected to the head end multi carrier modem for controllingtransmission of the downstream telephony information and downstreamcontrol data and for controlling receipt of the upstream control dataand upstream telephony information. The system includes a plurality ofservice units. Each service unit is associated with at least one remoteunit and operatively connected to the distribution network for upstreamtransmission of upstream telephony information and upstream control datain the second frequency bandwidth and for receipt of the downstreamcontrol data and downstream telephony information in the first frequencybandwidth. Each service unit includes a service unit multicarrier modemfor modulating at least the upstream telephony information on at leastone carrier orthogonal to at least one other carrier in the secondfrequency bandwidth and for demodulating at least the downstreamtelephony information modulated on at least a band of a plurality oforthogonal carriers in the first frequency bandwidth. Each service unitalso includes a service unit controller operatively connected to theservice unit multicarrier modem for controlling the modulation of anddemodulation performed by the service unit multicarrier modem. Theservice unit controller adjusts at least one local transmissioncharacteristic in response to an adjustment command from the head endcontroller transmitted in the downstream control data to the at leastone remote unit. The head end controller further includes a detector fordetecting the at least one local transmission characteristic of theservice unit modem associated with the at least one remote unit and forgenerating the adjustment command as a function of the detected at leastone transmission characteristic for transmittal to the service unitassociated with the at least one remote unit in the downstream controldata.

Distributed system control is also employed for acquiring and trackingremote service units newly connected to (or activated within) thesystem. An acquisition process makes rough estimates of the frequency,phase, and data-symbol timing of the head-end transmitter, calculatesthe round-trip delay of data to and from the head end, and then trackschanges in the frequency, phase, and timing of the head end, all withminimal overhead to the transmission of payload data within the system.A special non-valid data signal is used to signal the start of atraining pattern for acquisition purposes. Maintaining accurate powerbalancing or leveling among the remote units transmitting upstream tothe head end is both necessary and difficult in a multipoint-to-pointmulticarrier system. One embodiment of the invention uses both anautomatic gain control or equalizer at the head end and a transmitteroutput or power control at the remote end to achieve the conflictinggoals of wide dynamic range and high resolution amplitude control.

Furthermore, a communication system having a distribution networkbetween a head end and a plurality of remote units using a scanningmethod is described. The system includes the transmission, from the headend, of a plurality of modulated orthogonal carriers having telephonyinformation modulated thereon in a plurality of regions of a firstfrequency bandwidth. Each of the regions has at least one controlchannel associated therewith having control information modulatedthereon. A scanner at the remote units, scans each of the plurality ofregions in the first frequency bandwidth and locks onto the at least onecontrol channel associated with each of the plurality of regions todetect a unique identifier to determine which region of the firstfrequency bandwidth the remote unit is to tune to and which region in asecond frequency bandwidth the remote unit is to transmit within.

In another embodiment, the communication system includes a distributionnetwork between a head end and a plurality of remote units. The head endincludes a head end terminal for downstream transmission of downstreamcontrol data and downstream telephony information in a first frequencybandwidth over the distribution network and for receipt of upstreamtelephony information and upstream control data in a second frequencybandwidth over the distribution network. The head end terminal includesa head end multicarrier modem for modulating at least downstreamtelephony information on a plurality of orthogonal carriers in aplurality of regions of the first frequency bandwidth. The head endmulticarrier modem also demodulates at least upstream telephonyinformation modulated on a plurality of orthogonal carriers of aplurality of regions in the second frequency bandwidth. The plurality oforthogonal carriers in each of the regions includes a plurality oftelephony information channels for transmission of telephony informationthereon with each of the regions having at least one control channelassociated therewith for transmission of control data. The head endterminal also includes a head end controller operatively connected tothe head end multicarrier modem for controlling transmission of thedownstream telephony information and downstream control data and forcontrolling receipt of the upstream control data and upstream telephonyinformation. The system further includes a plurality of service unitmodems with each service unit modem associated with at least one remoteunit and operatively connected to the distribution network for upstreamtransmission of upstream telephony information and upstream control datain one of the plurality of regions of the second frequency bandwidth andfor receipt of the downstream control data and downstream telephonyinformation in one of the plurality of regions in the first frequencybandwidth. Each service unit modem includes a scanner for scanning eachof the plurality of regions in the first frequency bandwidth and forlocking onto the at least one control channel in each of the pluralityof regions to detect a unique identifier for each service unit modem todetermine which region of the first frequency bandwidth the service unitmodem is to tune to and which region in the second frequency bandwidththe service unit modem is to transmit within.

This invention further provides a three-part RAM structure, thefunctions of which can be permuted between input, conversion, and outputfunctions. In one embodiment, the conversion RAM section is configuredto offer four values to be accessed simultaneously in order to speedoperations.

In another embodiment, this invention relates to the field of electroniccommunication systems, and more specifically to an improved method andapparatus for providing a Sigma-Delta analog-to-digital conversion anddecimation for a modem.

According to another aspect of the invention there is provided a datadelivery system having a head end connected to a plurality of remotesubscribers over a network, the data originating from general purposedigital computers, wherein the data is transmitted to the remotesubscribers in a plurality of data channels, and at the start of a datatransmission session, one or more channels are selectively assigned tocarry the data between the head end and the subscriber premises witheach of the data channels maintaining a substantially constant rate ofdata transfer between the head end and the destination premise. Thesystem and method further allows that the number of assigned channelsassigned to a particular subscriber can be changed from one connectionto another to accommodate changes in overall system loading, but at alltimes maintaining a minimum number of assigned channels so that aminimum rate of data transfer can be maintained between the head end anda subscriber premise. The system also provides asymmetrical operation sothat the number of data channels assigned in the downstream path fromthe head end to the subscribers is much greater in number than thenumber of upstream data channels.

According to another aspect of the invention there is provided a systemof computer data and telephony data transmission over atelecommunications network having a head end connected to a plurality ofremote subscribers, the computer data originating from general purposedigital computers, and the telephony data received from or conveyed tothe public telephone network. The system comprises transmitting thecomputer data and telephony data to the remote subscribers in aplurality of data channels, and establishing a computer data ortelephony data connection between the head end and a subscriber premiseindependently of one another. Each of the computer data or telephonydata connections are established by assigning one or more of the datachannels to carry the computer data and one or more of the data channelsto carry the telephony data with at least some of the data channelsbeing available to carry either computer data or telephony data. Thesystem further allows that the number assigned data channels can bechanged from one connection to another to the same subscriber so thatthe overall bandwidth of the network can be reallocated. The systemfurther allows that the data channels are transmitted using RadioFrequency (RF) signaling, and that the RF is scrambled, therebyproviding data security.

According to yet another aspect of the invention, the system allows thata computer data received at the head end can initiate a connection to aremote subscriber, wherein the connection comprises the assignment ofone or more data channels to carry the computer data to the subscriber.

According to yet another embodiment, the system 10 or 500 of the presentinvention includes an ATM modulator which can receive ATM data andmodulate it onto the HFC network. In one preferred embodiment, digitalvideo data is delivered over an ATM network, multiplexed and modulatedonto the HFC in RF digital OFDM format on data connections establishedbetween the head end and a subscriber, as for example described abovewith respect to system 500. A digital set top box receives the digitalvideo, for example in 4.0 Mbps MPEG or equivalent, and converts it tovideo for display on a television. A return path over a telephony ordata channel allows for interactive digital video.

In another embodiment, a method for transmitting data over atelecommunication system from a head end to a service unit is provided.The service unit is assigned to subband of a transmission channel of thetelecommunication system. The subband includes a number of payloadchannels that transmit data at a first rate and a control channel thattransmits data at a second rate. The second rate is slower than thefirst rate. The system receives a request to transmit data to a serviceunit at the second, slower rate. The system further determines whetherto transmit the data at the first, faster, rate based on the size of thedata. When a payload channel in the subband is available to transmit thedata at the first rate, the system allocates the payload channel totransmit the data to the service unit over the payload channel at thefirst rate. When the payload channels are allocated to service units andat least one of the allocated payload channels is idle, the systemallocates the idle payload channel to transmit the data to the serviceunit over the payload channel at the first rate. This method can be usedto download software and transmission protocols as well as other datathat is not time sensitive. Further, the method advantageously providesflexibility in providing bandwidth for transmission of data in thetelecommunications system.

In another embodiment, a telecommunications system dynamically allocatesbandwidth among a plurality of service units. The system comprises ahead end that transmits data over a transmission medium to the serviceunits. The head end includes a modem circuit for narrow bandtransmission in at least one transmission channel. Each transmissionchannel includes a number of subbands having a number of payloadchannels and a control channel in each subband. Further, a controlcircuit in the head end assigns each service unit to a subband fortransmission and receipt of data. The control circuit also allocates apayload channel to a service unit in response to a request for bandwidthfor a service unit.

In another embodiment, the control circuit assigns a number of serviceunits to each subband. The control circuit dynamically allocatesbandwidth to the service units for selective use of the payload channelsin the subband. This increases the number of service units that can becoupled to the system.

In another embodiment, a method for dynamically allocating bandwidth toa service unit in a telecommunications system is provided. The systemuses a multichannel transmission scheme with transmission channels thatinclude a number of subbands. Each subband further includes a number ofpayload channels. The method begins by receiving a request for a payloadchannel for a service unit that is assigned to a first subband. Themethod selects an available payload channel in the first subband anddetermines if the payload channel is acceptable for providing service tothe service unit, for example, acceptable transmission quality. When thepayload channel is acceptable, the method allocates the payload channelto the service unit. When, however, the payload channel is unacceptable,the method selects other channels to find an acceptable payload channel.

In another embodiment, a telecommunications system implements a methodfor allocating payload channels for a service that use multiple payloadchannels to communicate with a service unit. The system assigns anidentifier for each payload channel that indicates the relative order ofthe multiple payload channels for the service. The system furthermonitors the quality of the payload channels of the system. When thequality of one of the multiple payload channels drops below a threshold,the system allocates a different payload channel to replace the originalpayload channel for the service. Once reallocated, the system uses theidentifier for the original payload channel so that the proper order forthe allocated payload channels is maintained by the service irrespectiveof the order that the payload channels are received at the service unit.

In another embodiment, a telecommunications system provides a method forusing an upstream payload channel to inform the head end of errors thatoccur in downstream payload channels. The system, monitors a downstreamtransmission channel at a service unit for transmission errors. Further,the system generates a signal at the service unit that indicatestransmission errors in the downstream payload channel. The system alsotransmits the signal to the head end on an associated upstream payloadchannel thus allowing the head end to monitor and respond to theperformance of the service unit and associated payload channels.

In another embodiment, a method for controlling a plurality of serviceunits in a telecommunication system is provided. The method firstassigns an identifier to each service unit. The method further assignseach service unit to a subband of a transmission channel of a narrowband transmission scheme. In the transmission channel, each subbandincludes a control channel for receiving and transmitting controlsignals. The method broadcasts the control signals for the service unitsover the control channels. The method identifies the terminal to use thecontrol signal with the identifier.

In another embodiment, a service unit for use with a communicationsystem transmits signals with a narrow band transmission scheme. Thetransmission channel is divided into a number of subbands with eachsubband including a number of payload channels and a control channel.The service unit includes a modem that is tunable to receive telephonyand control signals on a subband of a transmission channel. The serviceunit further includes a controller circuit coupled to the modem toreceive control signals over the control channel and to determine whichcontrol signals to use to control the operation of the modem. Theservice unit also includes interface circuits coupled to the controllerfor providing signals to a channel unit.

In another embodiment, a method for controlling power usage at a serviceunit of a telecommunications system is provided. The method comprisesdetermining the type of service supported by each line of a serviceunit. When the service unit supports analog telephony service, themethod determines the hook status of all of the lines of the serviceunit. When the lines are on-hook, the method powers down the serviceunits to conserve power usage until a request is received to use a lineof the service unit.

The present invention describes a method of establishing communicationbetween a head end and a plurality of remote units in a multi-point topoint communication system, such as when a fault as described above hasleft many users of the system without service. The method includestransmitting information from the head end to the plurality of remoteunits in a plurality of regions of a first frequency bandwidth. Each ofthe regions has at least one control channel associated therewith. Theinformation transmitted includes identification informationcorresponding to each of n remote units of the plurality of remoteunits. Such information is periodically transmitted for the n remoteunits from the head end on the at least one control channel of one ofthe plurality of regions of the first frequency bandwidth during a firstpredetermined time period. The identification information for each ofthe plurality of n remote units is transmitted out of phase with respectto the identification information for the other of the n remote units.At each of the n remote units, the at least one control channel of eachof the plurality of regions in the first frequency bandwidth is scannedto detect identification information corresponding to each of the nremote units to identify a particular region of the plurality of regionsthat each of the n remote units is to use for receiving information fromthe head end.

In one embodiment, a region is identified in a second frequencybandwidth in which each of the n remote units is to transmit within. Themethod further includes serially performing synchronization for each ofthe n remote units for communication with the head end, during a secondpredetermined time period after the first predetermined time period.

A multi-point to point communication system having a distributionnetwork between a head end and a plurality of remote units foraccomplishing the above method includes means for transmittinginformation from the head end to the plurality of remote units in aplurality of regions of a first frequency bandwidth. Each of the regionshas at least one control channel associated therewith. The transmittingmeans further periodically transmits identification informationcorresponding to each of a set of n remote units of the plurality ofremote units on at least one control channel of one of the plurality ofregions of the first frequency bandwidth during a first predeterminedtime period of an identification and synchronization time period. Theidentification information for each of the plurality of n remote unitsis transmitted out of phase with respect to the identificationinformation for the other of the n remote units. The system furtherincludes at each of the n remote units, means for scanning the at leastone control channel of each of the plurality of regions in the firstfrequency bandwidth to detect identification information during thefirst predetermined time period corresponding to each of the n remoteunits to identify a particular region of the plurality of regions thateach of the n remote units is to use for receiving information from thehead end. Further, at each of the n remote units, the system includesmeans for modulating at least upstream telephony information on at leastone carrier in a second frequency bandwidth orthogonal at the head endterminal to at least one other carrier in the second frequency bandwidthand for adjusting at least one local transmission characteristic inresponse to an adjustment command from the head end. Means at the headend for detecting the at least one local transmission characteristic ofeach of the n remote units and for generating the adjustment commands asa function of the detected at least one transmission characteristic fortransmittal to the n remote units to serially perform synchronizationfor each of the n remote units during a second predetermined time periodof the identification and synchronization time period is also includedin the system.

The present invention is a hybrid fiber/coax video and telephonycommunication network which integrates bi-directional telephony andinteractive video services into one network including optical fiber andcoaxial cable distribution systems. The present invention utilizesoptical fiber as the transmission medium for feeding a plurality ofoptical distribution nodes with video and telephony information from ahead end. Coaxial cable distribution systems are utilized for connectionof the distribution nodes to a plurality of remote units. The head endoptically transmits the video information downstream to the nodes whereit is converted to electrical signals for distribution to the remoteunits. Telephony information is also optically transmitted to the nodesin frequency bandwidths unused by the video information. The downstreamtelephony and video optical signals are converted to electricaltelephony and video signals for distribution to the plurality of remoteunits. The network provides for transmission of upstream electrical datasignals, for example telephony signals, to the head end by transmittingfrom the remote units upstream electrical data signals to thedistribution nodes where such upstream electrical data signals areconverted to upstream optical signals for transmission to the head end.

In one embodiment, the head end includes a first distribution terminalhaving at least one optical transmitter for transmitting opticaldownstream telephony signals on at least one optical fiber. In addition,the head end includes a second distribution terminal having a separateoptical transmitter for transmitting an optical downstream video signalon an optical fiber line.

In another embodiment, the video and telephony signal distributionnetwork transmits optical downstream video and telephony signals on atleast one optical fiber in a first frequency bandwidth. In thisembodiment, a second frequency bandwidth is reused for transmission ofupstream electrical data signals generated at the remote units. Thesecond frequency bandwidth is reused for transmission by each remoteunit.

In another embodiment of the invention, a filter is utilized at serviceunits which interface the coaxial distribution systems to userequipment. The ingress filter allows for passage of downstream videosignals to video equipment units and blocks downstream telephony signalstransmitted in a different frequency bandwidth.

DRAWINGS

FIG. 1 shows a block diagram of a communication system utilizing ahybrid fiber/coax distribution network;

FIG. 2 is an alternate embodiment of the system of FIG. 1;

FIG. 3 is a detailed block diagram of a host digital terminal (HDT) withassociated transmitters and receivers of the system of FIG. 1;

FIG. 4 is a block diagram of the associated transmitters and receiversof FIG. 3;

FIG. 5 is a block diagram of an optical distribution node of the systemof FIG. 1;

FIGS. 6, 7 are embodiments of frequency shifters for use in the opticaldistribution node of FIG. 5 and the telephony upstream receiver of FIG.4, respectively;

FIG. 8 is a general block diagram of an integrated service unit (ISU)such as a home integrated service unit (HISU) or a multiple integratedservice unit (MISU) of FIG. 1;

FIGS. 9, 10, 11 show data frame structures and frame signaling utilizedin the HDT of FIG. 3;

FIG. 12 is a general block diagram of a coax master card (CXMC) of acoax master unit (CXMU) of FIG. 3;

FIG. 13 shows a spectral allocation for a first transport embodiment fortelephony transport in the system of FIG. 1;

FIG. 14 shows a mapping diagram for QAM modulation;

FIG. 15 shows a mapping diagram for BPSK modulation;

FIG. 16 shows a subband diagram for the spectral allocation of FIG. 13;

FIGS. 17, 18 show alternative mapping diagrams or constellations for QAMmodulation;

FIG. 19 shows a timing diagram of an identification and synchronizationprocess;

FIG. 20 shows a timing diagram of a burst identification andsynchronization process;

FIG. 21 is a block diagram of a master coax card (MCC) downstreamtransmission architecture of the CXMU for the first transport embodimentof the system of FIG. 1;

FIG. 22 is a block diagram of a coax transport unit (CXTU) downstreamreceiver architecture of an MISU for the first transport embodiment ofthe system of FIG. 1;

FIG. 23 is a block diagram of a coax home module (CXHM) downstreamreceiver architecture of an HISU for the first transport embodiment ofthe of the system of FIG. 1;

FIG. 24 is a block diagram of a CXHM upstream transmission architectureassociated with the CXHM downstream receiver architecture of FIG. 23;

FIG. 25 is a block diagram of a CXTU upstream transmission architectureassociated with the CXTU downstream receiver architecture of FIG. 22;

FIG. 26 is a block diagram of an MCC upstream receiver architectureassociated with the MCC downstream transmission architecture of FIG. 21;

FIG. 27 is a flow diagram of a acquisition distributed loop routine foruse with the system of FIG. 1;

FIG. 28 is a flow diagram of a tracking distributed loop architectureroutine for use with the system of FIG. 1;

FIG. 29 shows a magnitude response of a polyphase filter bank of the MCCupstream receiver architecture of FIG. 26;

FIG. 30 is an enlarged view of part of the magnitude response of FIG.29;

FIG. 31 is a block diagram of an ingress filter structure and FFT of theMCC upstream receiver architecture of FIG. 26;

FIG. 32 is a block diagram of a polyphase filter structure of theingress filter structure and FFT of FIG. 31;

FIG. 33 is a block diagram of a carrier, amplitude, timing recoveryblock of the downstream receiver architectures of the first transportembodiment;

FIG. 34 is a block diagram of a carrier, amplitude, timing recoveryblock of the MCC upstream receiver architecture of the first transportembodiment;

FIG. 35 is a block diagram of internal equalizer operation for thereceiver architectures of the first transport embodiment;

FIG. 36 is a spectral allocation of a second transport embodiment fortransport in the system of FIG. 1;

FIG. 37 is a block diagram of an MCC modem architecture of the CXMU forthe second transport embodiment of the system of FIG. 1;

FIG. 38 is a block diagram of a subscriber modem architecture of theHISU for the second transport embodiment of the system of FIG. 1;

FIG. 39 is a block diagram of a modem of the subscriber modemarchitecture of FIG. 38;

FIG. 40 is a block diagram for channel monitoring used in the system ofFIG. 1;

FIGS. 41, 42, 43 are flow diagrams for error monitor portions of channelmonitor routines of FIG. 40;

FIG. 44 is an alternate flow diagram for the diagram of FIG. 42;

FIG. 45 is a flow diagram for a background monitor portion of thechannel monitor routines of FIG. 40;

FIG. 46 is a flow diagram for a backup portion of the channel monitorroutines of FIG. 40;

FIGS. 47, 48 are flow diagrams of an acquisition distributed looproutine for use with another embodiment of the system of FIG. 1;

FIG. 49 is a flow diagram of a downstream tracking loop for use with theembodiment of FIGS. 47 and 48;

FIG. 50 is a flow diagram of an upstream tracking loop for use with theembodiment of FIGS. 47 and 48;

FIG. 51 is a block diagram showing the locking of all clocks within asystem;

FIGS. 52, 53 depict phase diagrams of symbol waveforms;

FIGS. 54, 55, 56, 57 describe error rates and message-encoding methodsfor use in a system;

FIG. 58 is a block diagram of a scrambler;

FIG. 59 is a block diagram of a control circuit for a CXMU of an HDT ina telecommunications system;

FIGS. 60, 61, 62 are flow charts that illustrate methods for assigningsubbands and allocating payload channels in a telecommunications systemthat uses a multi-carrier communication scheme;

FIGS. 63, 64, 65, 66, 67 are frequency spectrum diagrams that illustrateexamples of assigning service units to subbands;

FIG. 68 is a flow chart that illustrates error monitoring by the channelmanager;

FIG. 69 is a flow chart that illustrates a method for allocating an ISUdata-link (IDL) channel in a telecommunications system;

FIG. 70 is a block diagram of FFT system 2100;

FIG. 71 is a block diagram of modem 2400 which includes a FFT system2100 configured to perform an IFFT in transmitter section 2401 andanother FFT system 2100 configured to perform an FFT in receiver section2402;

FIG. 72 is a block diagram of three logical banks of RAM: an input RAM2251, an output RAM 2253, and a conversion RAM 2252;

FIG. 73 is a block diagram of one embodiment of a physicalimplementation which provides the function of input RAM 2241, conversionRAM 2242, and output RAM 2243;

FIG. 74 is a block diagram of one embodiment of a dual radix core 2600;

FIGS. 75, 76, 77, 78, 79, 80, 81, 82 together form a table showing theorder of calculations for a “normal butterfly sub-operation”;

FIGS. 83, 84, 85, 86, 87, 88, 89, 90 together form a table showing theorder of calculations for a “transposed butterfly sub-operation”;

FIG. 91 is a block diagram of one embodiment of dual-radix core 2600showing the nomenclature used for the products output by multipliers2620 through 2627 and for addersubtractor-accumulators 2633;

FIG. 92 is a block diagram of one embodiment of anadder-subtractor-accumulator 2633;

FIG. 93 is a block diagram of modem 2400 which includes a Sigma-DeltaADC and decimator system to drive FFT system 2100;

FIG. 94 is a more detailed block diagram of modem receiver 2402;

FIG. 95 is a detailed block diagram of one embodiment of a Sigma-Deltaconverter 2840;

FIG. 96 is an overall schematic diagram of the data delivery transportsystem according to the present invention;

FIG. 97 is a simplified block diagram of the head-end terminal 12 of thesystem 500 according to the present invention;

FIG. 98 illustrates a Personal Cable Data Modem (PCDM) 540 and a DataModem Service Module (DMSM) 550;

FIG. 99 illustrates in greater detail a PCDM 540;

FIG. 100 illustrates a Data Modem Channel Unit (DMCU) 560;

FIG. 101 shows a graph of average bandwidth per user as a function ofthe number of users for the system 500;

FIG. 102 is a simplified block diagram of the data transport and framingof the system 500;

FIG. 103 illustrates a Local Area Network Unit (LANU) 580;

FIG. 104 illustrates in more detail a DMSM 550;

FIG. 105 illustrates in more detail a DMCU 560;

FIGS. 106, 107, 108, 109 illustrate the call setup for a data connectionon the system 500;

FIG. 110 illustrates a call termination sequence on the system 500;

FIG. 111 illustrates the software of a LANU 580;

FIG. 112 illustrates a PCDM 620 adapted for asymmetrical data delivery;

FIG. 113 illustrates the head-end configuration for asymmetrical datadelivery;

FIGS. 114, 115 illustrate another alternate embodiment of the inventionwherein digital video is received over an ATM network and transmittedover a modified form of system 10/500;

FIG. 116 shows a block diagram of a hybrid fiber/coax network;

FIG. 117 is a block diagram of a head end host distribution terminal ofthe network of FIG. 116;

FIG. 118 is a block diagram of an optical distribution node of thenetwork of FIG. 116;

FIG. 119 is a block diagram of a home coaxial line unit of the networkof FIG. 116;

FIG. 120 is a block diagram of an alternative embodiment fortransmission from the head end to the optical distribution nodes;

FIG. 121 is a block diagram of an impulse shaping technique utilized inaccordance with the present invention;

FIG. 122 is a block diagram of an alternative embodiment of the opticalto electrical converter of the head end host distribution terminal ofFIG. 117;

FIG. 123 is a block diagram of an alternative embodiment of the head endhost distribution terminal of FIG. 117; and

FIG. 124 is a flow chart that illustrates an embodiment of a method forcommunicating errors in a communication system.

DETAILED DESCRIPTION

The communication system 10, as shown in FIG. 1, is an access platformprimarily designed to deliver residential and business telecommunicationservices over a hybrid fiber-coaxial (HFC) distribution network 11. Thesystem 10 is a cost-effective platform for delivery of telephony andvideo services. Telephony services may include standard telephony,computer data and/or telemetry. In addition, the system 10 is a flexibleplatform for accommodating existing and emerging services forresidential subscribers.

The hybrid fiber-coaxial distribution network 11 utilizes optical fiberfeeder lines to deliver telephony and video service to a distributionnode 18 (referred to hereinafter as the optical distribution node (ODN))remotely located from a central office or a head end 32. From the ODNs18, service is distributed to subscribers via a coaxial network. Severaladvantages exist by utilizing the HFC-based communication system 10. Byutilizing fiber installed in the feeder, the system 10 spreads the costof optoelectronics across hundreds of subscribers. Instead of having aseparate copper loop which runs from a distribution point to eachsubscriber (“star” distribution approach), the system 10 implements abused approach where a distribution coaxial leg 30 passes each home andsubscribers “tap” the distribution coaxial leg 30 for service. Thesystem 10 also allows non-video services to be modulated fortransmission using more cost-effective RF modem devices in dedicatedportions of the RF spectrum. Finally, the system 10 allows videoservices to be carried on existing coaxial facilities with no additionalsubscriber equipment because the coaxial distribution links can directlydrive existing cable-ready television sets.

It should be apparent to one skilled in the art that the modem transportarchitecture described herein and the functionality of the architectureand operations surrounding such architecture could be utilized withdistribution networks other than hybrid fiber coax networks. Forexample, the functionality may be performed with respect to wirelesssystems. Therefore, the present invention contemplates use of suchsystems in accordance with the accompanying claims.

The system 10 includes host digital terminals 12 (HDTs) which implementall common equipment functions for telephony transport, such as networkinterface, synchronization, DSO grooming, and operations,administration, maintenance and provisioning (OAM&P) interfaces, andwhich include the interface between the switching network and atransport system which carries information to and from customerinterface equipment such as integrated service units 100 (ISUs), showngenerally in FIG. 8. Integrated services units (ISUs) 100, such as homeintegrated service units (HISUs) 68 or multiple user integrated serviceunits (MISUs) 66, which may include a business integrated service unitas opposed to a multiple dwelling integrated service unit, implement allcustomer interface functions and interface to the transport system whichcarries information to and from the switched network. In the presentsystem, the HDT 12 is normally located in a central office and the ISUs100 are remotely located in the field and distributed in variouslocations. The HDT 12 and ISUs 100 are connected via the hybridfiber-coax distribution network 11 in a multi-point to pointconfiguration. In the present system, the modem functionality requiredto transport information over the HFC distribution network 11 isperformed by interface equipment in both the HDT 12 and the ISUs 100.Such modem functionality is performed utilizing orthogonal frequencydivision multiplexing.

The communication system shall now be generally described with referenceto FIGS. 1, 3 and 8. The primary components of system 10 are hostdigital terminals (HDTs) 12, video host distribution terminal (VHDT) 34,telephony downstream transmitter 14, telephony upstream receiver 16, thehybrid fiber coax (HFC) distribution network 11 including opticaldistribution node 18, and integrated service units 66, 68 (showngenerally as ISU 100 in FIG. 8) associated with remote units 46. The HDT12 provides telephony interface between the switching network (notedgenerally by trunk line 20) and the modem interface to the HFCdistribution network for transport of telephony information. Thetelephony downstream transmitter 14 performs electrical to opticalconversion of coaxial RF downstream telephony information outputs 22 ofan HDT 12, shown in FIG. 3, and transmits onto redundant downstreamoptical feeder lines 24. The telephony upstream receiver 16 performsoptical to electrical conversion of optical signals on redundantupstream optical feeder lines 26 and applies electrical signals oncoaxial RF upstream telephony information inputs 28 of HDT 12. Theoptical distribution node (ODN) 18 provides interface between theoptical feeder lines 24 and 26 and coaxial distribution legs 30. The ODN18 combines downstream video and telephony onto coaxial distributionlegs 30. The integrated services units provide modem interface to thecoaxial distribution network and service interface to customers.

The HDT 12 and ISUs 100 implement the telephony transport systemmodulator-demodulator (modem) functionality. The HDT 12 includes atleast one RF MCC modem 82, shown in FIG. 3 and each ISU 100 includes anRF ISU modem 101, shown in FIG. 8. The MCC modems 82 and ISU modems 101use a multi-carrier RF transmission technique to transport telephonyinformation, such as DSO+ channels, between the HDT 12 and ISUs 100.This multi-carrier technique is based on orthogonal frequency divisionmultiplexing (OFDM) where a bandwidth of the system is divided up intomultiple carriers, each of which may represent an information channel.Multi-carrier modulation can be viewed as a technique which takestime-division multiplexed information data and transforms it tofrequency-division multiplexed data. The generation and modulation ofdata on multiple carriers is accomplished digitally, using an orthogonaltransformation on each data channel. The receiver performs the inversetransformation on segments of the sampled waveform to demodulate thedata. The multiple carriers overlap spectrally. However, as aconsequence of the orthogonality of the transformation, the data in eachcarrier can be demodulated with negligible interference from the othercarriers, thus reducing interference between data signals transported.Multi-carrier transmission obtains efficient utilization of thetransmission bandwidth, particularly necessary in the upstreamcommunication of a multi-point to point system. Multi-carrier modulationalso provides an efficient means to access multiple multiplexed datastreams and allows any portion of the band to be accessed to extractsuch multiplexed information, provides superior noise immunity toimpulse noise as a consequence of having relatively long symbol times,and also provides an effective means for eliminating narrowbandinterference by identifying carriers which are degraded and inhibitingthe use of these carriers for data transmission (such channel monitoringand protection is described in detail below). Essentially, the telephonytransport system can disable use of carriers which have interference andpoor performance and only use carriers which meet transmission qualitytargets.

Further, the ODNs 18 combine downstream video with the telephonyinformation for transmission onto coaxial distribution legs 30. Thevideo information from existing video services, generally shown by trunkline 20, is received by and processed by head end 32. Head end 32 or thecentral office, includes a video host distribution terminal 34 (VHDT)for video data interface. The VHDT 34 has optical transmittersassociated therewith for communicating the video information to theremote units 46 via the ODNs 18 of the HFC distribution network II.

The telephony transmitter 14 of the HDTs 12, shown in FIGS. 3 and 4,includes two transmitters for downstream telephony transmission toprotect the telephony data transmitted. These transmitters areconventional and relatively inexpensive narrow band laser transmitters.One transmitter is in standby if the other is functioning properly. Upondetection of a fault in the operating transmitter, the transmission isswitched to the standby transmitter. In contrast, the transmitter of theVHDT 34 is relatively expensive as compared to the transmitters of HDT12 as it is a broad band analog DFB laser transmitter. Therefore,protection of the video information, a non-essential service unliketelephony data, is left unprotected. By splitting the telephony datatransmission from the video data transmission, protection for thetelephony data alone can be achieved. If the video data information andthe telephony data were transmitted over one optical fiber line by anexpensive broad band analog laser, economies may dictate that protectionfor telephony services may not be possible. Therefore, separation ofsuch transmission is of importance.

Further with reference to FIG. 1, the video information is opticallytransmitted downstream via optical fiber line 40 to splitter 38 whichsplits the optical video signals for transmission on a plurality ofoptical fiber lines 42 to a plurality of optical distribution nodes 18.The telephony transmitter 14 associated with the HDT 12 transmitsoptical telephony signals via optical fiber feeder line 42 to theoptical distribution nodes 18. The optical distribution nodes 18 convertthe optical video signals and optical telephony signals for transmissionas electrical outputs via the coaxial distribution portion of the hybridfiber coax (HFC) distribution network 11 to a plurality of remote units46. The electrical downstream video and telephony signals aredistributed to ISUs via a plurality of coaxial legs 30 and coaxial taps44 of the coaxial distribution portion of the HFC distribution network11.

The remote units 46 have associated therewith an ISU 100, showngenerally in FIG. 8, that includes means for transmitting upstreamelectrical data signals including telephony information, such as fromtelephones and data terminals, and in addition may include means fortransmitting set top box information from set top boxes 45 as describedfurther below. The upstream electrical data signals are provided by aplurality of ISUs 100 to an optical distribution node 18 connectedthereto via the coaxial portion of the HFC distribution network 11. Theoptical distribution node 18 converts the upstream electrical datasignals to an upstream optical data signal for transmission over anoptical fiber feeder line 26 to the head end 32.

FIG. 2 generally shows an alternate embodiment for providingtransmission of optical video and optical telephony signals to theoptical distribution nodes 18 from head end 32, the HDT 12 and VHDT 34in this embodiment utilize the same optical transmitter and the sameoptical fiber feeder line 36. The signals from HDT 12 and VHDT 34 arecombined and transmitted optically from head end 32 to splitter 38. Thecombined signal is then split by splitter 38 and four split signals areprovided to the optical distribution nodes 18 for distribution to theremote units by the coaxial distribution legs 30 and coaxial taps 44.Return optical telephony signals from the ODNs 18 would be combined atsplitter 38 for provision to the head end. However, as described above,the optical transmitter utilized would be relatively expensive due toits broad band capabilities, lessening the probabilities of being ableto afford protection for essential telephony services.

As one skilled in the art will recognize, the fiber feeder lines 24, 26,as shown in FIG. 1, may include four fibers, two for transmissiondownstream from downstream telephony transmitter 14 and two fortransmission upstream to upstream telephony receiver 16. With the use ofdirectional couplers, the number of such fibers may be cut in half. Inaddition, the number of protection transmitters and fibers utilized mayvary as known to one skilled in the art and any listed number is notlimiting to the present invention as described in the accompanyingclaims.

The present invention shall now be described in further detail. Thefirst part of the description shall primarily deal with video transport.The remainder of the description shall primarily be with regard totelephony transport.

Video Transport

The communication system 10 includes the head end 32 which receivesvideo and telephony information from video and telephony serviceproviders via trunk line 20. Head end 32 includes a plurality of HDTs 12and a VHDT 34. The HDT 12 includes a network interface for communicatingtelephony information, such as T1, ISDN, or other data servicesinformation, to and from telephony service providers, such communicationalso shown generally by trunk line 20. The VHDT 34 includes a videonetwork interface for communicating video information, such as cable TVvideo information and interactive data of subscribers to and from videoservice providers, such communication also shown generally by trunk line20.

The VHDT 34 transmits downstream optical signals to a splitter 38 viavideo optical fiber feeder line 40. The passive optical splitter 38effectively makes four copies of the downstream high bandwidth opticalvideo signals. The duplicated downstream optical video signals aredistributed to the correspondingly connected optical distribution nodes18. One skilled in the art will readily recognize that although fourcopies of the downstream video signals are created, any number of copiesmay be made by an appropriate splitter and that the present invention isnot limited to any specific number.

The splitter is a passive means for splitting broad band optical signalswithout the need to employ expensive broad band optical to electricalconversion hardware. Optical signal splitters are commonly known to oneskilled in the art and available from numerous fiber optic componentmanufacturers such as Gould, Inc. In the alternative, active splittersmay also be utilized. In addition, a cascaded chain of passive or activesplitters would further multiply the number of duplicated opticalsignals for application to an additional number of optical distributionnodes and therefore increase further the remote units serviceable by asingle head end. Such alternatives are contemplated in accordance withthe present invention as described by the accompanying claims.

The VHDT 34 can be located in a central office, cable TV head end, or aremote site and broadcast up to about 112 NTSC channels. The VHDT 34includes a transmission system like that of a LITEAMP™ system availablefrom American Lightwave Systems, Inc., currently a subsidiary of theassignee hereof. Video signals are transmitted optically by amplitudemodulation of a 1300 nanometer laser source at the same frequency atwhich the signals are received (that is, the optical transmission is aterahertz optical carrier which is modulated with the RF video signals).The downstream video transmission bandwidth is about 54-725 MHz. Oneadvantage in using the same frequency for optical transmission of thevideo signal as the frequency of the video signals when received is toprovide high bandwidth transmission with reduced conversion expense.This same-frequency transmission approach means that the modulationdownstream requires optical to electrical conversion or proportionalconversion with a photo diode and perhaps amplification, but nofrequency conversion. In addition, there is no sample data bandwidthreduction and little loss of resolution.

An optical distribution node 18, shown in further detail in FIG. 5,receives the split downstream optical video signal from the splitter 38on optical fiber feeder line 42. The downstream optical video signal isapplied to a downstream video receiver 400 of the optical distributionnode 18. The optical video receiver 400 utilized is like that availablein the LITEAMP product line. The converted signal from video receiver400, proportionally converted utilizing photodiodes, is applied tobridger amplifier 403 along with converted telephony signals fromdownstream telephony receiver 402. The bridger amplifier 403simultaneously applies four downstream electrical telephony and videosignals to diplex filters 406 which allow for full duplex operation byseparating the transmit and receive functions when signals of twodifferent frequency bandwidths are utilized for upstream and downstreamtransmission. There is no frequency conversion performed at the ODN 18with respect to the video or the downstream telephony signals as thesignals are passed through the ODNs to the remote units via the coaxialportion of the HFC distribution network 11 in the same frequencybandwidth as they are received at the ODNs 18.

After the ODN 18 has received the downstream optical video signals andsuch signals are converted to downstream electrical video signals, thefour outputs of the ODN 18 are applied to four coaxial legs 30 of thecoaxial portion of the HFC distribution network 11 for transmission ofthe downstream electrical video signals to the remote units 46. Suchtransmission for the electrical video signals occurs in about the 54-725MHz bandwidth. Each ODN 18 provides for the transmission on a pluralityof coaxial legs 30 and any number of outputs is contemplated inaccordance with the present invention as described in the accompanyingclaims.

As shown in FIG. 1, each coaxial cable leg 30 can provide a significantnumber of remote units 46 with downstream electrical video and telephonysignals through a plurality of coaxial taps 44. Coaxial taps arecommonly known to one skilled in the art and act as passivebidirectional pickoffs of electrical signals. Each coaxial cable leg 30may have a number of coaxial taps 44 connected in series. In addition,the coaxial portion of the HFC distribution network 11 may use anynumber of amplifiers to extend the distance data can be sent over thecoaxial portion of such HFC distribution network 11.

Downstream video signals are provided from the coaxial taps 44 to theremote units 46. The video signal from the coaxial tap 44 is provided toan HISU 68 which is generally shown by the block diagram of ISU 100 inFIG. 8. The ISU 100 is provided with the downstream electrical video andtelephony signal from tap 44 and it is applied to diplex filter 104. Thedownstream electrical video and telephony signal is passed through thediplex filter 104 to both an ingress filter 105 and ISU modem 101. Thedownstream video signal is passed by the ingress filter 105 to videoequipment via an optional set top box 45. The downstream electricaltelephony signal applied from the diplex filter 104 to the ISU modem 101is processed as described in further detail below.

Ingress filter 105 provides the remote unit 46 with protection againstinterference of signals applied to the video equipment as opposed tothose provided to other user equipment such as telephones or computerterminals. Ingress filter 105 passes the video signals; however, itblocks those frequencies not utilized by the video equipment. Byblocking those frequencies not used by the video equipment, straysignals are eliminated that may interfere with the other services by thenetwork to at least the same remote unit.

The set top box 45 is an optional element at the remote unit 46.Interactive video data from set top box 45 would be transmitted by anadditional separate RF modem provided by the video service provider at arelatively low frequency in the bandwidth of about 5 to 40 MHz. Suchfrequency must not be one used for the transport of upstream anddownstream telephony data and downstream video.

For an MISU 66, a separate coaxial line from coaxial tap 44 is utilizedto provide transmission of video signals from the coaxial tap 44 to theset top box 45 and thus for providing downstream video signals to videoequipment 47. The ingress filter 105 as shown in FIG. 8 is not a part ofthe MISU 66 as indicated by its dashed representation.

Alternative embodiments of the VHDT 34 may employ other modulation andmixing schemes or techniques to shift the video signals in frequency,and other encoding methods to transmit the information in a codedformat. Such techniques and schemes for transmitting analog video data,in addition to those transmitting digital video data, are known to oneskilled in the art and are contemplated in accordance with the spiritand scope of the present invention as described in the accompanyingclaims.

Telephony Transport

With reference to FIG. 3, telephony information and ISU operations andcontrol data (hereinafter referred to as control data) modulated oncarriers by MCC modem 82 is transmitted between the HDT 12 and thetelephony downstream transmitter 14 via coaxial lines 22. Telephonyinformation and control data modulated on carriers by ISUs 100 isreceived at telephony upstream receiver 16 and communicated to the MCCmodem 82 via coaxial cable lines 28. The telephony downstreamtransmitter 14 and the telephony upstream receiver 16 transmit andreceive, respectively, telephony information and control data viaoptical fiber feeder lines 24 and 26 to and from a corresponding opticaldistribution node 18. The control data may include all operations,administration, maintenance & provisioning (OAM&P) for providing thetelephony services of the system 10 and any other control data necessaryfor providing transport of telephony information between the HDT 12 andthe ISUs 100.

A block diagram of the HDT 12 is shown in FIG. 3. The HDT 12 includesthe following modules: Eight DS1 Units (DSIU) (seven quad-DS1 units 48plus one protection unit 50), one protection switch & test conversionunit 52 (PSTU), two clock & time slot interchange units 54 (CTSUs) (oneactive and one standby/protection unit), six coax master units 56(CXMUs) (three active and three standby/protection units), two shelfcontrol units 58 (SCNUs) (one active and one standby/protection unit),and two power supply units 60 (PWRUs) (two load-sharing units whichprovide the appropriate HDT voltages from a central office supply). TheDS1U units can also be adapted to transfer data in the standard E1Uformat, if desired.

The HDT 12 comprises all the common equipment functions of the telephonytransport of the communication system 10. The HDT 12 is normally locatedin a central office and directly interfaces to a local digital switch ordigital network element equipment. The HDT provides the networkinterface 62 for all telephony information. Each HDT accommodates from 2to 28 DSX-1 inputs at the network interface 62, representing a maximumof 672 DSO channels. The HDT 12 also provides all synchronization fortelephony transport in the system 10. The HDT 12 may operate in anyoneof three synchronization modes: external timing, line timing or internaltiming. External timing refers to synchronization to a buildingintegrated timing supply reference which is sourced from a centraloffice in which the HDT 12 is located. Line timing is synchronized tothe recovered clock from a DSX-1 signal normally derived from the localdigital switch. Internal timing is a free-running or hold-over operationwhere the HDT maintains its own synchronization in the absence of anyvalid reference inputs.

The HDT 12 also provides quarter-DSO grooming capabilities andimplements a 4096×4096 full-access, non-blocking quarter-DSO (16 kbps)cross-connect capability. This allows DSOs and quarter-DSOs (ISDN “D”channels) to be routed from any times lot at the DSX-1 network interface62 to any customer serviced by any ISU 100.

The HDT 12 further provides the RF modem functionality required fortelephony transport over the HFC distribution network 11 including theMCC modem 82. The HDT 12 accommodates up to three active CXMUs 56 forproviding the modem interface to the HFC distribution network 11 andalso provides one-for-one protection for each active CXMU 56.

The HDT 12 coordinates the telephony transport system including controland communication of many ISUs of the multi-point to point communicationsystem 10. Each HDT 12 module performs a function. The DS1U module 48provides the interface to the digital network and DSX-1 termination. ThePSTU 52 provides DS1U equipment protection by switching the protectionDS1U 50 for a failed DS1U module 48. The CTSU 54 provides thequarter-DSO times lot grooming capability and all system synchronizationfunctions. The CTSU 54 also coordinates all call processing in thesystem. The CXMU 56, described in further detail below, provides themodem functionality and interface for the OFDM telephony transport overthe HFC distribution network 11 and the SCNU 58 supervises the operationof the entire communication system providing all OAM&P functions fortelephony transport. Most processing of requests for provisioning isperformed by the SCNU 58.

Downstream Telephony Transmitter

The downstream telephony transmitter 14, shown in FIG. 4, takes thecoaxial RF outputs 22 from the active CXMUs 56 of the HDT 12 which carrytelephony information and control data and combines the outputs 22 intoa downstream telephony transmission signal. The electrical-to-opticalconversion logic required for the optical transmission is implemented ina stand-alone downstream telephony transmitter 14 rather than in the HDT12 to provide a more cost effective transport solution. By placing thisfunction in a separate component, the expense of this function does notneed to be replicated in each CXMU 56 of the HDT 12. This reduces thecost of the CXMU 56 function and allows the CXMU 56 to transmit andreceive over coax instead of fiber. The downstream telephony transmitter14 also provides for transmission on redundant downstream fiber feederlines 24 to an ODN 18.

The downstream telephony transmitter 14 is co-located with the HDT 12preferably within a distance of 100 feet or less. The downstreamtelephony transmitter 14 receives the coaxial RF outputs from the activeCXMUs 56, each within a 6 MHz frequency band, and combines them atcombiner 25 into a single RF signal. Each 6 MHz frequency band isseparated by a guard band as is known to one skilled in the art.Downstream telephony information is then transmitted in about the725-800 MHz frequency band. The telephony transmitter 14 passes thecombined signal through a 1-to-2 splitter (not shown), thereby producingredundant downstream electrical signals. The two redundant signals areeach delivered to redundant laser transmitters 501 forelectrical-to-optical conversion and the redundant signals modulate anoptical output such that the output of the downstream telephonytransmitter 14 is on two optical feeder lines 24, each having anidentical signal modulated thereon. This provides protection for thedownstream telephony portion of the present system. Both Fabry-Perotlasers in the telephony transmitter 14 are active at all times. Allprotection functions are provided at the receive end of the opticaltransmission (located at the ODN 18) where one of two receivers isselected as “active”; therefore, the telephony transmitter 14 requiresno protection switching capabilities.

Upstream Telephony Receiver

The upstream telephony receiver 16 performs the optical-to-electricalconversion on the upstream optical telephony signals on the upstreamoptical feeder lines 26 from the ODN 18. The upstream telephony receiver16 is normally co-located in the central office with the HDT 12, andprovides an electrical coaxial output to the HDT 12, and a coaxialoutput 23 to be provided to a video set-top controller (not shown).Upstream telephony information is routed via coax lines 28 from theupstream telephony receiver 16 to active CXMUs 56 of the HDT 12. Thecoaxial link 28 between the HDT 12 and the upstream telephony receiver16 is preferably limited to a distance of 100 feet or less and is anintra-office link. Video set-top controller information, as described inthe Video Transport section hereof, is located in a bandwidth of the RFspectrum of 5-40 MHz which is not utilized for upstream telephonytransport such that it is transmitted along with the upstream telephonyinformation.

The upstream telephony receiver 16 has dual receivers 502 for the dualupstream optical fiber feeders lines 26. These feeder lines 26 carryredundant signals from the ODN 18 which contain both telephonyinformation and control data and also video set-top box information. Theupstream telephony receiver 16 performs automatic protection switchingon the upstream feeder lines 26 from the ODN. The receiver 502 selectedas “active” by protection logic is split to feed the coaxial outputs 28which drive the HDT 12 and output 23 is provided to the set-topcontroller (not shown).

Optical Distribution Node

Referring to FIG. 5, the ODN 18 provides the interface between theoptical feeder lines 24 and 26 from the HDT 12 and the coaxial portionof the HFC distribution network 11 to the remote units 46. As such, theODN 18 is essentially an optical-to-electrical and electrical-to-opticalconverter. The maximum distance over coax of any ISU 100 from an ODN 18is preferably about 6 km and the maximum length of the combined opticalfeeder line/coaxial drop is preferably about 20 km. The optical feederline side of the ODN 18 terminates six fibers although such number mayvary. They include: a downstream video feeder line 42 (single fiber fromvideo splitter 38), a downstream telephony feeder line 24 (fromdownstream telephony transmitter 14), a downstream telephony protectionfeeder line 24 (from downstream telephony transmitter 14), an upstreamtelephony feeder line 26 (to upstream telephony receiver 16), anupstream protection feeder line 26 (to upstream telephony receiver 16),and a spare fiber (not shown). The ODN 18 provides protection switchingfunctionality on the receive optical feeder lines 24 from the downstreamtelephony transmitter. The ODN provides redundant transmission on theupstream optical feeder lines 26 to the upstream telephony receiver.Protection on the upstream optical feeder lines is controlled at theupstream telephony receiver 16. On the coaxial distribution side of ODN18, the ODN 18 terminates up to four coaxial legs 30.

In the downstream direction, the ODN 18 includes downstream telephonyreceiver 402 for converting the optical downstream telephony signal intoan electrical signal and a bridger amplifier 403 that combines it withthe converted downstream video signal from downstream video receiver 400terminated at the ODN 18 from the VHDT 34. This combined wide-bandelectrical telephony/video signal is then transported in the spectrumallocated for downstream transmission, for example, the 725-800 MHzband, on each of the four coaxial legs of the coaxial portion of the HFCdistribution network 11. As such, this electrical telephony and videosignal is carried over the coaxial legs 30 to the ISUs 100; the bridgeramplifier 403 simultaneously applying four downstream electricaltelephony and video signals to diplex filters 406. The diplex filters406 allow for full duplex operation by separating the transmit andreceive functions when signals at two different frequency bandwidths areutilized for upstream and downstream transmission. There is no frequencyconversion available at the ODN 18 for downstream transport as thetelephony and video signals are passed through the ODN 18 to the remoteunits 46 via the coaxial portion of HFC distribution network 11 in thesame frequency bandwidth as they are received at the ODN 18. As shown inFIG. 1, each coaxial leg 30 can provide a significant number of remoteunits 46 with downstream electrical video and telephony signals througha plurality of coaxial taps 44. Coaxial taps 44 commonly known to oneskilled in the art act as passive bidirectional pickoffs of electricalsignals. Each coaxial leg 30 may have a number of coaxial taps connectedin a series. In addition, the coaxial portion of the HFC distributionnetwork 11 may use any number of amplifiers to extend the distance datacan be sent over the coaxial portions of the system 10. The downstreamelectrical video and telephony signals are then provided to an ISU 100(FIG. 8), which, more specifically, may be an HISU 68 or an MISU 66 asshown in FIG. 1.

In the upstream direction, telephony and set top box information isreceived by the ODN 18 at diplex filters 406 over the four coaxial legs30 in the RF spectrum region from 5 to 40 MHz. The ODN 18 may includeoptional frequency shifters 64 equipped on up to three of four coaxiallegs 30. These frequency shifters 64, if utilized, mix the upstreamspectrum on a coaxial leg to a higher frequency prior to combining withthe other three coaxial legs. Frequency shifters 64 are designed toshift the upstream spectrum in multiples of 50 MHz. For example, thefrequency shifters 64 may be provisioned to mix the upstream informationin the 5-40 MHz portion of the RF spectrum to any of the followingranges: 50 to 100 MHz, 100 to 150 MHz, or 150 to 200 MHz. This allowsany coaxial leg 30 to use the same portion of the upstream RF spectrumas another leg without any spectrum contention when the upstreaminformation is combined at the ODN 18. Provisioning of frequencyshifters is optional on a coaxial leg 30. The ODN 18 includes combiner408 which combines the electrical upstream telephony and set top boxinformation from all the coaxial legs 30 (which mayor may not befrequency shifted) to form one composite upstream signal having allupstream information present on each of the four coaxial legs 30. Thecomposite electrical upstream signal is passively 1:2 split and eachsignal feeds an upstream Fabry-Perot laser transmitter which drives acorresponding upstream fiber feeder line 26 for transmission to theupstream telephony receiver 16.

FIG. 7 illustrates an embodiment of a frequency shifter, indicatedgenerally at 64′, for use in ODN 18 of FIG. 5. Frequency shifter 64′comprises a mixer 700 that is coupled to receive and shift the frequencyband of RF signals in the upstream direction from diplex filter 406 fora coaxial leg 30. An output of mixer 700 is coupled through a bandpassfilter 704 to combiner 408. Local oscillator 702 is coupled to provide asignal to control the operation of mixer 700.

In operation, frequency shifter 64′ shifts a block of RF signals from afirst frequency range to a second frequency range. For example, asmentioned above, the RF signals provided to frequency shifter 64′ maycomprise RF signals in the range from 5 to 40 MHz. In one embodiment,ODN 18 comprises three frequency shifters 64′. In this embodiment, thelocal oscillators 702 of the three frequency shifters provide signals of76 MHz, 149 MHz, and 222 MHz, respectively. Thus, frequency shifters 64′respectively shift the upstream RF signals approximately to the 50 to100 MHZ, 125 to 175 MHZ and 200 to 250 MHZ ranges.

If the upstream telephony and set top box signals are upshifted at theODN 18, the upstream telephony receiver 16 includes frequency shifters31 to downshift the signals according to the upshifting done at the ODN18. A combiner 33 then combines the downshifted signals for applicationof a combined signal to the HDT 12. Such downshifting and combining isonly utilized if the signals are upshifted at the ODN 18.

FIG. 6 illustrates an embodiment of a frequency shifter, indicatedgenerally at 31′, for use in telephony upstream receiver 16 of FIG. 3.Frequency shifter 31′ returns a block of RF signals shifted by frequencyshifter 64′ to original frequency range of the block. For example,frequency shifter 31′ may return a block of RF signals to 5 to 40 MHzfrom 50 to 100 MHz.

As discussed in more detail below, the upstream telephony signalsprocessed by frequency shifters 31′ and 64′ are typically OFDM signals.Thus, frequency shifters 64′ must return the RF signals to the originalfrequency range without introducing adverse phase and frequency errors.To reduce the likelihood of this corruption of the OFDM signals,frequency shifter 31′ locks its local oscillator to the local oscillatorof a corresponding frequency shifter 64′ using a pilot tone transmittedfrom ODN 18 to telephony upstream receiver 16.

Frequency shifter 31′ includes a bandpass filter 706 that is coupled toreceive an RF signal from ODN 18. Bandpass filter 706 is coupled to asplitter 708. Splitter 708 is coupled to provide the RF signal to aninput of mixer 718. Further, splitter 708 provides a second output thatis used to generate a local oscillator signal for mixer 718. This localoscillator signal is phase locked with a corresponding local oscillator702 of frequency converter 64′. This second output of splitter 708 iscoupled to phase detector 712 through bandpass filter 710. Phasedetector 712 is coupled to provide a control signal to voltagecontrolled oscillator 714. Voltage controlled oscillator 714 is coupledthrough splitter 716 to provide the local oscillator signal to mixer718. Splitter 716 further provides a feedback signal to phase detector712.

In operation, phase detector 712 phase locks local oscillator signal offrequency shifter 31′ with local oscillator 702 of a correspondingfrequency shifter 64′. Phase detector 712 compares the pilot tone fromODN 18 with the feedback signal from voltage controlled oscillator 714to generate the control signal for voltage controlled oscillator 714.Consequently, the local oscillator signal provided to mixer 718 is phaselocked with the corresponding local oscillator 702 of frequency shifter64′. Mixer 718 uses the local oscillator signal from splitter 716 andvoltage controlled oscillator 714 to shift the block of RF signalsreceived by frequency shifter 31′ to the original frequency range of theblock of RF signals. Advantageously, unacceptable modifications of theOFDM upstream signal by frequency shifters 64′ and 31′ are thus avoided.

Integrated Services Unit (ISUs)

Referring to FIG. 1, the ISUs 100, such as HISU 68 and MISU 66, providethe interface between the HFC distribution network 11 and the customerservices for remote units 46. Two basic types of ISUs are shown, whichprovide service to specific customers. Multiple user integrated serviceunit 66 (MISUs) may be a multiple dwelling integrated service unit or abusiness integrated service unit. The multiple dwelling integratedservice unit may be used for mixed residential and businessenvironments, such as multi-tenant buildings, small businesses andclusters of homes. These customers require services such as plain oldtelephone service (POTS), data services, DSI services, and standardTR-57 services. Business integrated service units are designed toservice business environments. They may require more services, forexample, data services, ISDN, DSI services, higher bandwidth services,such as video conferencing, etc. Home integrated services units 68(HISUs) are used for residential environments such as single-tenantbuildings and duplexes, where the intended services are POTS and basicrate integrated digital services network (ISDN). Description for ISUsshall be limited to the HISUs and MISUs for simplicity purposes asmultiple dwelling and business integrated service units have similarfunctionality as far as the present invention is concerned.

All ISUs 100 implement RF modem functionality and can be genericallyshown by ISU 100 of FIG. 8. ISU 100 includes ISU modem 101, coax slavecontroller unit (CXSU) 102, channel units 103 for providing customerservice interface, and diplex filter/tap 104. In the downstreamdirection, the electrical downstream telephony and video signal isapplied to diplex filter/tap 104 which passes telephony information toISU modem 101 and video information to video equipment via an ingressfilter 105 in the case of a HISU. When the ISU 100 is a MISU 66, thevideo information is rejected by the diplex filter. The ISU modem 101demodulates the downstream telephony information utilizing a modemcorresponding to the MCC modem 82 used for modulating such informationon orthogonal multicarriers at HDT 12. ISU 100 demodulates downstreamtelephony information from a coaxial distribution leg 30 in aprovisionable 6 MHz frequency band. Timing generation 107 of the ISUmodem 101 provides clocking for CXSU 102 which provides processing andcontrols reception and transmission by ISU modem 101. The demodulateddata from ISU modem 101 is passed to the applicable channel units 103via CXSU 102 depending upon the service provided. For example, thechannel units 103 may include line cards for POTS, DS1 services, ISDN,other data services, etc. Each ISU 100 provides access to a fixed subsetof all channels available in a 6 MHz frequency band corresponding to oneof the CXMUs of HDT 12. This subset of channels varies depending uponthe type of ISU 100. An MISU 66 may provide access to many DSO channelsin a 6 MHz frequency band, while an HISU 68 may only provide access to afew DSO channels.

The channel units 103 provide telephony information and control data tothe CXSU 102, which provides such data to ISU modem 101 and controls ISUmodem 101 for modulation of such telephony data and control data in aprovisional 6 MHz frequency band for transmission onto the coaxialdistribution leg 30 connected thereto. The upstream 6 MHz frequency bandprovisionable for transmission by the ISU 100 to the HDT 12 correspondsto one of the downstream 6 MHz bands utilized for transmission by theCXMUs 56 of HDT 12.

The CXSU 102 which applies demodulated data from the ISU modem 101 tothe applicable channel units, performs data integrity checking on thedownstream 10 bit DSO+ packets received from the ISU modem 101. Each tenbit DSO+ packet as described below includes a parity or data integritybit. The CXSU 102 will check the parity of each downstream 10 bit DSO+channel it receives. Further, the parity of each upstream DSO+ receivedfrom the channel units 103 is calculated and a parity bit inserted asthe tenth bit of the upstream DSO+ for decoding and identification bythe HDT 12 of an error in the upstream data. If an error is detected byCXSU 102 when checking the parity of a downstream 10 bit DSO+ channel itreceives, the parity bit of the corresponding upstream channel will beintentionally inverted to inform the HDT 12 of a parity error in thedownstream direction. Therefore, the upstream parity bit is indicativeof errors in the downstream DSO+ channel and the corresponding upstreamDSO+ channel. An example of such a parity bit generation process isdescribed in U.S. patent application Ser. No. 08/074,913 (now U.S. Pat.No. 5,519,830) entitled “Point-to Multipoint Performance Monitoring andFailure Isolation System” assigned to the assignee hereof and entirelyincorporated herein by reference. This upstream parity bit is utilizedin channel monitoring as described further below. As would be apparentto one skilled in the art, the parity checking and generation may beperformed, at least in part, in other elements of the ISU or associatedtherewith such as the channel units.

Each ISU 100 recovers synchronization from downstream transmission,generates all clocks required for ISU data transport and locks theseclocks to the associated HDT timing. The ISUs 100 also provide callprocessing functionality necessary to detect customer line seizure andline idle conditions and transmit these indications to the HDT 12. ISUs100 terminate and receive control data from the HDT 12 and process thecontrol data received therefrom. Included in this processing aremessages to coordinate dynamic channel allocation in the communicationsystem 10. Finally, ISUs 100 generate ISU operating voltages from apower signal received over the HFC distribution network 11 as shown bythe power signal 109 taken from diplex filter/tap 104.

Data Path in HDT

The following is a detailed discussion of the data path in the hostdigital terminal (HDT) 12. Referring to FIG. 3, the data path betweenthe network facility at the network interface 62 and the downstreamtelephony transmitter 14 proceeds through the DS1U 48, CTSU 54, and CXMU56 modules of the HDT 12, respectively, in the downstream direction.Each DS1U 48 in the HDT 12 takes four DS1s from the network and formatsthis information into four 24-channel, 2.56 Mbps data streams ofmodified DSO signals referred to as CTSU inputs 76. Each DSO in the CTSUinput has been modified by appending a ninth bit which can carrymultiframe timing, signaling information and control/status messages(FIG. 9). This modified DSO is referred to as a “DSO+.” The ninth bitsignal (NBS) carries a pattern which is updated each frame and repeatsevery 24 frames. This maps each 64 kbps DSO from the network into a 72kbps DSO+. Thus, the twenty-four DSO channels available on each DS1 areformatted along with overhead information into twenty-four DSO+ channelson each of four CTSU input streams.

The ninth bit signaling (NBS) is a mechanism developed to carry themultiframe timing, out-of-band signaling bits and miscellaneous statusand control information associated with each DSO between the DS1U andthe channel units. Its main functions are to carry the signaling bits tochannel units 103 and to provide a multiframe clock to the channel units103 so that they can insert upstream bit signaling into the DSO in thecorrect frame of the multiframe. Because downstream DSOs may be comingfrom DS1s which do not share the same multiframe phase each DSO mustcarry a multiframe clock or marker which indicates the signaling framesassociated with the origination DS1. The NBS provides this capability.Ninth bit signaling is transparent to the OFDM modem transport of thecommunication system 10.

Up to eight DS1Us 48 may be equipped in a single HDT 12; including sevenactive DS1Us 48 and a protection DS1U module 50. Thus, 32 CTSU inputsare connected between the DS1Us and the CTSUs 54 but a maximum of 28 canbe enabled to carry traffic at any one time. The four remaining CTSUinputs are from either the protection DS1U or a failed DS1U. The PSTUincludes switch control for switching the protection DS1U 50 for afailed DS1U.

Each CTSU input is capable of carrying up to 32, 10-bit channels, thefirst 24 channels carry DSO+s and the remaining bandwidth is unused.Each CTSU input 76 is clocked at 2.56 Mbps and is synchronized to the 8kHz internal frame signal (FIG. 11). This corresponds to 320 bits per125 us frame period. These 320 bits are framed as shown in FIG. 9. Thefourteen gap bits 72 at the beginning of the frame carry only a singleactivity pulse in the 2nd bit position, the remaining 13 bits are notused. Of the following 288 bits, the first 216 bits normally carrytwenty-four DSO+ channels where each DSO+ corresponds to a standard 64kbps DSO channel plus the additional 8 kbps signaling bit. Thus, eachDSO+ has a bandwidth of 72 kbps (nine bits every 8 kHz frame). Theremaining 72 bits are reserved for additional DSO+ payload channels. Thefinal eighteen bits 74 of the frame are unused gap bits.

The clock and time slot interchange unit 54 (CTSU) of the HDT 12 takesinformation from up to 28 active CTSU input data streams 76 andcross-connects them to up to twenty-four 32-channel, 2.56 Mbps outputdata streams 78 which are input to the coax master units (CXMUs) 56 ofthe HDT 12. The format of the data streams between the CTSU 54 and theCXMUs 56 is referred to as a CTSU output. Each CTSU output can alsocarry up to 32, 10-bit channels like the CTSU input. The first 28 carrytraffic and the remaining bandwidth is unused. Each CTSU output isclocked at 2.56 Mbps and is synchronized to the 8 kHz internal framingsignal of the HDT 12 (FIG. 11). This corresponds to 320 bits per 125 psframe period. The frame structure for the 320 bits are as describedabove for the CTSU input structure.

The HDT 12 has the capability of time and space manipulation ofquarter-DSO packets (16 kbps). This function is implemented with thetime slot interchange logic that is part of CTSU 54. The CTSU implementsa 4096×4096 quarter-DSO cross-connect function, although not all timeslots are utilized. In normal operation, the CTSU 54 combines andrelocates up to 672 downstream DSO+ packets (or up to 2688 quarter-DSOpackets) arranged as 28 CTSU inputs of 24 DSO+s each, into 720 DSO+packets (or 2880 quarter-DSO packets) arranged as 24 CTSU outputs of 32DSO+s each.

The system has a maximum throughput of 672 DSO+ packets at the networkinterface so not all of the CTSU output bandwidth is usable. If morethan the 672 channels are assigned on the “CTSU output” side of theCTSU, this implies concentration is being utilized. Concentration isdiscussed further below.

Each CXMU 56 is connected to receive eight active CTSU outputs 78 fromthe active CTSU 54. The eight CTSU outputs are clocked by a 2.56 MHzsystem clock and each carries up to 32 DSO+s as described above. TheDSO+s are further processed by the CXMU 56 and a tenth parity bit isappended to each DSO+ resulting in a 10 bit DSO+. These 10 bit packetscontain the DSO, the NBS (ninth bit signal) and the parity or dataintegrity bit (FIG. 10). The 10 bit packets are the data transmitted onthe HFC distribution network 11 to the ISUs 100. The 10th bit or dataintegrity bit inserted in the downstream channels is decoded and checkedat the ISU and utilized to calculate and generate a parity bit forcorresponding channels in the upstream as described above. This upstreamparity bit which may be representative of an error in the downstream orupstream channel is utilized to provide channel protection or monitoringas further described herein.

CTSU 54 is also capable of applying a conventional Reed-Soloman code totransmitted payload data for running error correction. Such codes carryan overhead which must be borne by the inclusion of error-correctingsymbols in each block or message transmitted. Generally, two checksymbols are required to correct one corrupted data symbol in a message.(The incorrect symbol may contain any number of errors in its 5 bits, aslong as all bit errors are confined to the same symbol. But even asingle incorrect bit in each of two symbols counts as two errors.) Shortmessages impose less computational overhead on a system, but can correctfewer errors in the message. Conversely, long messages require morecomputation and more latency before the corrections can be applied, buttheir error-correction ability is greater. FIG. 54 represents, for anexample system, the probability of an uncorrectable error in a frame forvarious error probabilities in one individual symbol. The solid curveshows the error performance for a 21-frame message having 19 frames ofdata symbols and two frames of error-correction code; the dashed curverepresents a 41-frame message having 37 data and four code frames; thedotted curve gives the best performance, with 73 data frames and eightcode frames in an 81-frame message.

The present system allows a choice of different error-correctionabilities for different types of data. For example, voice data is highlyredundant, and needs little defense against errors. Financialtransaction data, on the other hand, wants a large degree of dataintegrity. In addition, it may be desirable to allow a user to select,and pay for, whatever degree of error correction that he desires. CTSU54, FIG. 3, includes a conventional “provisioning table,” whichspecifies a number of parameters relating to particular payloadchannels. FIG. 55 shows a provisioning table 4411 having an added columncontaining indications for several different amounts of errorprotection. In method 4410, step 4412 reads the entry for a particularchannel to be set up. In this implementation, the entry may specifymessage lengths of 21, 41, or 81 bits, respectively having the abilityto correct 1, 2, or 4 symbols; the entry may also specify no correction,in which case message blocks do not apply. Step 4413 encodes the tableentry in an IOC message and sends it to the ISU whose address appears inthat row of table 4411. A general-purpose processor in CXSU 102 of theISU stores the frame length in step 4414. As the CXSU receives data frommodem 101, FIG. 8, it decodes the frames of an entire message, 4415,then decodes the check symbols for the message, 4416, and signals anerror, 4417, if one exists in the message. Steps 4415-4417 repeat forsubsequent messages. The ISU employs the same process to send framesupstream to the head end, using the frame length setting specified instep 4414.

Within both CXSU 102 at the ISU and the CXMU, FIG. 12 at the HDT, a21-frame message or block requires 19 symbol or frame times to decodethe message, then has two frames of latency while its two check symbolsare decoded. A 41-frame message uses four frames of time for computationof any errors from the four check symbols following its 37 data symbols.An 81-frame message presents any error indication 8 frame times afterthe end of its 73 data frames. (One extra frame of delay is imposed inthe downstream direction due to remapping at the HDT.) If all messageswere to start at the same time for all channels in an entire band, thecomputational load in the HDT would peak during the check-symbol frames,and would be lower at other times. Since the processor must be capableof handling the peak loads, its power is underutilized at other times.

The present system allows a lower-capacity processor to handle errorcorrection by staggering the beginning times of different messages indifferent subbands of channels, so that not all of them come due at thesame time. That is, the start of a message in any channel of a subbandis offset from the start of a message or “multiframe” signal, to bedescribed, by a predetermined number of frame times of 125 ps each. Thetable below shows how the 24 subbands of FIG. 16 are offset, for eachmessage length which can be selected.

TABLE 1 Subband 21 frames 41 frames 81 frames 0 0 0 0 1 0 0 1 2 0 1 2 30 1 3 4 1 2 4 5 1 2 5 6 1 3 6 7 1 3 7 8 2 4 8 9 2 4 9 10 2 5 10 11 2 511 12 3 6 12

Only 13 subband settings are required, since no unit tunes more than 130channels. Giving all 10 channels of each subband the same offset doesnot overload the processors of the remote units. However, the head end(HDT), which receives and transmits all channels, can enjoy significantrelief from not having to encode or decode the check symbols for allchannels at the same time.

FIG. 56 shows steps 4420 for performing frame staggering. Step 4421repeats method 4420 for all active payload channels. Step 4422 accessesthe current messages for the channels in one subband. Step 4423calculates the 1, 2, or 4 Reed-Soloman check words for the 21, 41, or 81message data words. Step 4424 waits N frames past the start of amultiframe, whereupon step 4425 sends the message to modem 82, FIG. 3for transmission. At a remote ISU, CXSU performs the same steps 4420 forupstream messages. Step 4422 is required only in an MISU, because allchannels in an HISU reside in the same subband. Method 4420 may beperformed at both ends of the system, as described herein; it may alsobe performed only at one end, either HDT or ISU. Staggering from the ISUto the HDT is preferable if only one end is staggered, because the mostcritical processing load is the error-correction of all 240 channels inthe upstream receiving modem, FIG. 26.

The use of error-correcting codes along with unencoded data raisesproblems in a real-time transport system. Data arrives from the trunkline 20, FIG. 1, at a constant rate. This data must be transmitteddownstream in the same time duration, whether it is encoded along theway, or sent unencoded. Likewise, upstream data usually must betransmitted at the same rate whether or not it is encoded. That is, theuse of error-correcting codes must be time-transparent at both ends ofthe system. But error-correcting codes require the transmission of checkdigits or symbols along with the data. The present system resolves thisdifficulty by packing the data words differently if they are encoded. Asexplained above, the basic unencoded word length for a DSO++ channel isten bits: eight data bits, a signaling (NBS) bit, and a parity bit. Whenencoding is used, however, this format is changed to nine-bit words,with a single parity bit for the entire message. This is the reason forthe choice of frame sizes for the encoded modes. A 21 frame messagecontains 19 data frames, which would ordinarily be transmitted as10×19=190 bits. Those same data frames, packaged as nine-bit words alongwith two nine-bit check words, require (19+2)×9=189 bits; adding onemore parity bit covering the entire message lock gives 190 bits, thesame number as that required for the unencoded version of the same data.The 41-frame message has 37 frames of data, or 370 bits in unencoded10-bit format. Encoded as 37 nine-bit words along with four check words,the same message requires (37+4)×9=369 bits; again, a single additionalparity bit yield the 370 bits of the same data in unencoded form. The81-frame format has 73 data words, 8 check words, and a parity bit,yielding the same number of bits as 73 data words in 10-bit form.

There are many other combinations of numbers which yield similarresults. These can be found heuristically without a great deal ofexperimentation. The first step is to estimate rough numbers of large(parity-bearing) words in one or more message sizes, and the number oferrors desired to be correctable for each size. The next step is todetermine a number of smaller (non-parity) words that carry the sameamount of data, but which form a total message the same size or slightlysmaller than the total number of bits in the large-word format. Anyexcess bits then are assigned to parity over the block—or to any otherfunction, for that matter. For example, if two bits are left overinstead of one for each message, they could represent two parity bitsover the message, two control or format-designation bits, etc. The useof check symbols, of course, greatly reduces the need for parity orother forms of error detection. In fact, while the present system usesthe message-parity bit as parity in the downstream direction, the ISUdeliberately sets the parity bit to an incorrect value in an upstreammessage if it was incorrect in the downstream message. This serves tosignal the HDT that a bit error was encountered, when the HDT would nototherwise be aware of it; this in turn allows the HDT to keep moreaccurate statistics on channel quality for reallocating channels, or forother purposes.

FIG. 57 shows a method 4430 for adding the “code packing” feature to themethod 4420 of FIG. 56. Step 4431 repeats the steps for all channels.Step 4432 determines whether the data for the channel is to be encodedor not. If not, step 4433 merely transmits it word-by-word to the modem.If it is to be encoded, step 4434 strips the parity (or other) bites)from each word. After step 4435 has formed the check words, step 4436calculates the message-wide parity, or other desired function.Thereafter, step 4437 waits the proper number of frame times (asspecified by method 4420, and step 4338 sends the message to the modemas before. In the upstream direction, the reverse path through the HDTis substantially a mirror of the forward path through the HDT 12. Forexample, the tenth parity bit is processed at the CXMU 56 and the signalfrom the CXMU 56 to the CTSU 54 is in the format of FIG. 9.

The round trip delay of a DSO is the same for every data path. The timedelay over the path from the downstream CTSU output, through CXMU 56,over the HFC distribution network to the ISU 100 and then from the ISU100, back over the HFC distribution network 11, through CXMU 56 and toCTSU 54 is controlled by upstream synchronization, as described indetail below. Generally, path delay is measured for each ISU and if itis not the correct number of frames long, the delay length is adjustedby adding delay to the path at the ISU 100.

Coax Master Unit (CXMU)

The coax master unit 56 (CXMU), shown in FIG. 3, includes the coaxmaster card logic 80 (CXMC) and the master coax card (MCC) modem 82. Aspreviously described, up to six CXMUs may be equipped in an HDT 12. The6 CXMUs 56 include three pairs of CXMUs 56 with each pair providing fortransmit in a 6 MHz bandwidth. Each pair of CXMUs 56 includes one activeCXMU and a standby CXMD. Thus, one to one protection for each CXMU isprovided. As shown in FIG. 3, both CXMUs of the pair are provided withupstream telephony data from the upstream telephony receiver 16 and arecapable of transmitting via the coaxial line 22 to the downstreamtelephony transmitter 14. As such, only a control signal is required toprovide for the one-to-one protection indicating which CXMU 56 of thepair is to be used for transmission or reception.

Coax Master Card Logic (CXMC)

The coax master card logic 80 (CXMC) of the CXMU 56 (FIG. 12), providesthe interface between the data signals of the HDT 12, in particular ofthe CTSU 54, and the modem interface for transport of data over the HFCdistribution network 11. The CXMC 80 interfaces directly to the MCCmodem 82. The CXMC 80 also implements an ISU operations channeltransceiver for multi-point to point operation between the HDT 12 andall ISUs 100 serviced in the 6 MHz bandwidth in which the CXMU 56controls transport of data within. Referring to FIG. 12, the CXMCincludes controller and logic 84, downstream data conversion 88,upstream data conversion 90, data integrity 92, IOC transceiver 96, andtiming generator 94.

Downstream data conversion 88 performs the conversion from the nine-bitchannel format from CTSU 54 (FIG. 9) to the ten-bit channel format (FIG.10) and generates the data integrity bit in each downstream channeltransported over the HFC distribution network 11. The data integrity bitrepresents odd parity. Downstream data conversion 88 is comprised of atleast a FIFO buffer used to remove the 32 gap bits 72, 74 (FIG. 9)present in the downstream CTSU outputs and insert the tenth, dataintegrity bit, on each channel under control of controller and logic 84.

The upstream data conversion 90 includes at least a FIFO buffer whichevaluates the tenth bit (data integrity) appended to each of theupstream channels and passes this information to the data integritycircuitry 92. The upstream data conversion 90 converts the data streamof ten-bit channels (FIG. 10) back to the nine-bit channel format (FIG.9) for application to CTSU 54. Such conversion is performed undercontrol of controller and logic 84.

The controller and logic 84 also manages call processing and channelallocation for the telephony transport over the HFC distribution network11 and maintains traffic statistics over the HFC distribution network 11in modes where dynamic time-slot allocation is utilized, such as forproviding TR-303 services, concentration services commonly known tothose skilled in the art. In addition, the controller 84 maintains errorstatistics for the channels in the 6 MHz band in which the CXMUtransports data, provides software protocol for all ISU operationschannel communications, and provides control for the corresponding MCCmodem 82.

The data integrity 92 circuitry processes the output of the tenth bitevaluation of each upstream channel by the upstream conversion circuit90. In the present system, parity is only guaranteed to be valid on aprovisioned channel which has a call in progress. Because initializedand activated ISU transmitters may be powered down when the ISUs areidle, the parity evaluation performed by the CXMC is not always valid. Aparity error detected indicates either a transmission error in anupstream channel or a transmission error in a downstream channelcorresponding to the upstream channel.

The ISU operations channel (IOC) transceiver 96 of the CXMC 80 containstransmit buffers to hold messages or control data from the controllerand logic 84 and loads these IOC control messages which are a fixedtotal of 8 bytes in length into a 64 kbps channel to be provided to theMCC modem 82 for transport on the HFC distribution network 11. In thisimplementation, all IOC channels carry the same information at alltimes. That is, the IOC messages are broadcast simultaneously over allthe channels. This allows the use of inexpensive and rugged narrow-bandmodems in the ISUs, reserving the more expensive and critical widebandmodels for the HDT, which uses only one modem for an entire 6 MHz band,and which can be located centrally in a controlled environment. In theupstream direction, the IOC transceiver receives the 64 kbps channel viathe MCC modem 82 which provides the controller and logic 84 with suchmessages.

The timing generator circuit 94 receives redundant system clock inputsfrom both the active and protection CTSUs 54 of the HDT 12. Such clocksinclude a 2 kHz HFC multiframe signal, which is generated by the CTSU 54to synchronize the round trip delay on all the coaxial legs of the HFCdistribution network. This signal indicates multiframe alignment on theISU operations channel and is used to synchronize symbol timing and datareconstruction for the transport system. A 8 kHz frame signal isprovided for indicating the first “gap” bit of a 2.56 MHz, 32 channelsignal from the CTSU 54 to the CXMU 56. A 2.048 MHz clock is generatedby the CTSU 54 to the SCNU 58 and the CXMU 56. The CXMU 56 uses thisclock for ISU operations channel and modem communication between theCXMC 80 and the MCC modem 82. A 2.56 MHz bit clock is used for transferof data signals between the DS1Us 48 and CTSUs 54 and the CTSUs 54 andCXMCs 56. A 20.48 MHz bit clock is utilized for transfer of the 10-bitdata channels between the CXMC and the MCC.

Master Coax Card (MCC) Modem

The master coax card (MCC) modem 82 of the CXMU 56 interfaces on oneside to the CXMC 80 and on the other side to the telephony transmitter14 and receiver 16 for transmission on and reception from the HFCdistribution network 11. The MCC modem 82 implements the modemfunctionality for OFDM transport of telephony data and control data. Theblock diagram of FIG. 3 identifies the associated interconnects of theMCC modem 82 for both upstream and downstream communication. The MCCmodem 82 is not an independent module in the HDT 12, as it has nointerface to the HDT 12 other than through the CXMC 80 of the CXMU 56.The MCC modem 82 represents the transport system logic of the HDT 12. Assuch, it is responsible for implementing all requirements associatedwith information transport over the HFC distribution network 11. EachMCC modem 82 of the CXMUs 56 of HDT 12 is allocated a maximum bandwidthof 6 MHz in the downstream spectrum for telephony data and control datatransport. The exact location of the 6 MHz band is provisionable by theCXMC 80 over the communication interface via the IOC transceiver 96between the CXMC 80 and MCC modem 82. The downstream transmission oftelephony and control data is in the RF spectrum of about 725 to 800MHz.

Each MCC modem 82 is allocated a maximum of 6 MHz in the upstreamspectrum for receipt of control data and telephony data from the ISUswithin the RF spectrum of about 5 to 40 MHz. Again, the exact locationof the 6 MHz band is provisionable by the CXMC 80 over the communicationinterface between the CXMC 80 and the MCC modem 82.

The MCC modem 82 receives 256 DSO+ channels from the CXMC 80 in the formof a 20.48 MHz signal as described previously above. The MCC modem 82transmits this information to all the ISUs 100 using the multicarriermodulation technique based on OFDM as previously discussed herein. TheMCC modem 82 also recovers 256 DSO+ multicarrier channels in theupstream transmission over the HFC distribution network and convertsthis information into a 20.48 Mbps stream which is passed to CXMC 80. Asdescribed previously, the multicarrier modulation technique involvesencoding the telephony and control data, such as by quadrature amplitudemodulation, into symbols, and then performing an inverse FFT techniqueto modulate the telephony and control data on a set of orthogonalmulticarriers.

Symbol alignment is a necessary requirement for the multi carriermodulation technique implemented by the MCC modem 82 and the ISU modems101 in the ISUs 100. In the downstream direction of transmission, allinformation at an ISU 100 is generated by a single CXMU 56, so thesymbols modulated on each multicarrier are automatically phase aligned.However, upstream symbol alignment at a receiver of the MCC modem 82varies due to the multi-point to point nature of the HFC distributionnetwork 11 and the unequal delay paths of the ISUs 100. In order tomaximize receiver efficiency at the MCC modem 82, all upstream symbolsmust be aligned within a narrow phase margin. This is done by utilizingan adjustable delay parameter in each ISU 100 such that the symbolperiods of all channels received upstream from the different ISUs 100are aligned at the point they reach the HDT 12. This is part of theupstream synchronization process and shall be described further below.In addition, to maintain orthogonality of the multicarriers, the carrierfrequencies used for the upstream transmission by the ISUs 100 must befrequency locked to the HDT 12.

Incoming downstream information from the CXMC 80 to the MCC modem 82 isframe aligned to the 2 kHz and 8 kHz clocks provided to the MCC modem82. The 2 kHz multi-frame signal is used by the MCC modem 82 to conveydownstream symbol timing to the ISUs as described in further detailbelow. This multiframe clock conveys the channel correspondence andindicates the multi-carrier frame structure so that the telephony datamay be correctly reassembled at the ISU 100. Two kHz represents thegreatest common factor between 10 kHz (the modem symbol rate) and 8 kHz(the data frame rate).

All ISUs 100 will use the synchronization information inserted by theassociated MCC modem 82 to recover all downstream timing required by theISUs 100. This synchronization allows the ISUs 100 to demodulate thedownstream information and modulate the upstream transmission in such away that all ISU 100 transmissions received at the HDT 12 aresynchronized to the same reference. Thus, the carrier frequencies usedfor all ISU 100 upstream transmission will be frequency locked to theHDT 12.

The symbol alignment is performed over synchronization channels in thedownstream and upstream 6 MHz bandwidths under the responsibility of theMCC modem 82, in addition to providing path delay adjustment,initialization and activation, and provisioning over suchsynchronization channels until initialization and activation is completeas further described herein. These parameters are then tracked by use ofthe IOC channels. Because of their importance in the system, the IOCchannel and synchronization channels may use a different modulationscheme for transport of control data between the MCC modem 82 and ISUs100 which is more robust or of lesser order (less bits/sec/Hz orbits/symbol) than used for transport of telephony data. For example, thetelephony data may be modulated using quadrature amplitude modulation,while the IOC channel and synchronization channel may be modulatedutilizing BPSK modulation techniques. The MCC modem 82 also demodulatestelephony and control data modulated on multicarriers by the ISUs 100.Such demodulation is described further below with respect to the variousembodiments of the telephony transport system.

Functions with respect to the OFDM transport system for which the MCCmodem 82 is responsible, include at least the following, which arefurther described with respect to the various embodiments in furtherdetail. The MCC modem 82 detects a received amplitude/level of asynchronization pulse/pattern from an ISU 100 within a synchronizationchannel and passes an indication of this level to the CXMC 80 over thecommunication interface therebetween. The CXMC 80 then provides acommand to the MCC modem 82 for transmission to the ISU 100 beingleveled for adjustment of the amplitude level thereof. The MCC modem 82also provides for symbol alignment of all the upstream multicarriers bycorrelating an upstream pattern modulated on a synchronization channelwith respect to a known symbol boundary and passing a required symboldelay correction to the CXMC 80 over the communication interfacetherebetween. The CXMC 80 then transmits via the MCC modem 82 a messagedownstream to the ISU 100 to adjust the symbol delay of the ISU 100.

Likewise, with regard to synchronizing an ISU 100 for overall path delayadjustment, the MCC modem 82 correlates an upstream multiframe patternmodulated in the proper bandwidth by the ISU 100 on the IOC channel withrespect to a known reference boundary, and passes a required path delaycorrection to the CXMC 80 over the modem interface therebetween. TheCXMC 80 then transmits via the MCC modem 82 over the IOC channel amessage downstream to adjust the overall path delay of an ISU 100.

Summary of Bidirectional Multi-Point to Point Telephony Transport

The following summarizes the transport of telephony and controlinformation over the HFC distribution network 11. Each CXMU 56 of HDT 12is provisioned with respect to its specific upstream and downstreamoperating frequencies. The bandwidth of both upstream and downstreamtransmission by the CXMU 56 are a maximum of 6 MHz, with the downstreamtransmission in a 6 MHz band of the RF spectrum of about 725-800 MHz.

In the downstream direction, each MCC modem 82 of the CXMU 56 provideselectrical telephony and control data signals to the downstreamtelephony transmitter 14 via coaxial line 22 in its provisional 6 MHzbandwidth. The RF electrical telephony and control data signals from theMCC modems 82 of the HDT 12 are combined into a composite signal. Thedownstream telephony transmitter then passes the combined electricalsignal to redundant electrical-to-optical converters for modulation ontoa pair of protected downstream optical feeder lines 24.

The downstream optical feeder lines 24 carry the telephony informationand control data to an ODN 18. At the ODN 18, the optical signal isconverted back to electrical and combined with the downstream videoinformation (from the video head end feeder line 42) into an electricaldownstream RF output signal. The electrical RF output signal includingthe telephony information and control data is then fed to the fourcoaxial distribution legs 30 by ODN 18. All telephony information andcontrol data downstream is broadcast on each coaxial leg 30 and carriedover the coaxial portion of the HFC distribution network 11. Theelectrical downstream output RF signal is tapped from the coax andterminated on the receiver modem 101 of an ISU 100 through diplex filter104, shown in FIG. 8.

The RF electrical output signals include telephony information andcontrol data modulated on orthogonal multicarriers by MCC modem 82utilizing orthogonal frequency division multiplexing techniques; thetelephony information and control data being mapped into symbol data andthe symbols being modulated on a plurality of orthogonal carriers usingFFT techniques. As the symbols are all modulated on carriers at a singlepoint to be transmitted to multiple points in the system 10,orthogonality of the multicarriers and symbol alignment of the symbolsmodulated on the orthogonal multicarriers are automatically aligned fortransport over the HFC distribution network 11 and the telephonyinformation and control data is demodulated at the ISUs 100 by the modem101. The ISU 100 receives the RF signal tapped from the coax of thecoaxial portion of the HFC network 11. The RF modem 101 of the ISU 100demodulates the signal and passes the telephony information and controldata extracted to the CXSU controller 102 for provision to channel units103 as appropriate. The ISU 100 represents the interface where thetelephony information is converted for use by a subscriber or customer.

The CXMUs 56 of the HDT 12 and the ISUs 100 implement the bidirectionalmulti-point to point telephony transport system of the communicationsystem 10. The CXMUs 56 and the ISUs, therefore, carry out the modemfunctionality. The transport system in accordance with the presentinvention may utilize three different modems to implement the modemfunctionality for the transport system. The first modem is the MCC modem82 which is located in each CXMU 56 of the HDT 12. The HDT 12, forexample, includes three active MCC modems 82 (FIG. 3) and is capable ofsupporting many ISUs 100, representing a multi-point to point transportnetwork. The MCC modem 82 coordinates telephony information transport aswell as control data transport for controlling the ISUs 100 by the HDT12. For example, the control data may include call processing messages,dynamic allocation and assignment messages, ISU synchronization controlmessages, ISU modem control messages, channel unit provisioning, and anyother ISU operation, administration, maintenance and provisioning(OAM&P) information. The second modem is a single family subscriber orHISU modem optimized to support a single dwelling residential unit.Therefore, it must be low in cost and low in power consumption. Thethird modem is the multiple subscriber or MISU modem, which is requiredto generally support both residential and business services.

The HISU modem and the MISU modem may take several forms. For example,the HISU modem and the MISU modem may, as described further in detailbelow with regard to the various embodiments of the present invention,extract only a small portion of the multicarriers transmitted from theHDT 12 or a larger portion of the multicarriers transmitted from the HDT12. For example, the HISU may extract 20 multicarriers or 10 payloadchannels of telephony information transported from the HDT 12 and theMISU may extract information from 260 multicarriers or 130 payloadchannels transported from the HDT 12. Each of these modems may use aseparate receiver portion for extracting the control data from thesignal transported by the HDT 12 and an additional receiver portion ofthe HISU modem to extract the telephony information modulated on themulticarriers transported from the HDT 12. This shall be referred tohereinafter as an out of band ISU modem. The MCC modem 82 for use withan out of band ISU modem may modulate control information within theorthogonal carrier waveform or on carriers somewhat offset from suchorthogonal carriers. In contrast to the out of band ISU modem, the HISUand MISU modems may utilize a single receiver for the ISU modem andextract both the telephony information and control data utilizing thesingle receiver of the modem. This shall be referred to hereinafter asan in-band ISU modem. In such a case, the control data is modulated oncarriers within the orthogonal carrier waveform but may utilizedifferent carrier modulation techniques. For example, BPSK formodulation of control data on the carriers as opposed to modulation oftelephony data on payload carriers by QAM techniques. In addition,different modulation techniques may be used for upstream or downstreamtransmission for both control data and telephony data. For example,downstream telephony data may be modulated on the carriers utilizing 256QAM and upstream telephony data may be modulated on the carriersutilizing 32 QAM. Whatever modulation technique is utilized fortransmission dictates what demodulation approach would be used at thereceiving end of the transport system. Demodulation of the downstreamtelephony information and control data transported by the HDT 12 shallbe explained in further detail below with reference to block diagrams ofdifferent modem embodiments.

In the upstream direction, each ISU modem 101 at an ISU 100 transmitsupstream on at least one orthogonal multicarrier in a 6 MHz bandwidth inthe RF spectrum of about 5 to 40 MHz; the upstream 6 MHz bandcorresponding to the downstream 6 MHz band in which transmissions arereceived. The upstream electrical telephony and control data signals aretransported by the ISU modems 101 to the respectively connected opticaldistribution node 18 as shown in FIG. 1 via the individual coaxial cablelegs 30. At the ODN 18, the upstream signals from the various ISUs arecombined and transmitted optically to the HDT 12 via optical feederlines 26. As previously discussed, the upstream electrical signals fromthe various ISUs may, in part, be frequency shifted prior to beingcombined into a composite upstream optical signal. In such a case, thetelephony receiver 16 would include corresponding downshiftingcircuitry.

Due to the multi-point to point nature of transport over the HFCdistribution network 11 from multiple ISUs 100 to a single HDT 12, inorder to utilize orthogonal frequency division multiplexing techniques,symbols modulated on each carrier by the ISUs 100 must be aligned withina certain phase margin. In addition, as discussed in further detailbelow, the round trip path delay from the network interface 62 of theHDT 12 to all ISUs 100 and back from the ISUs 100 to the networkinterface 62 in the communication system 10 must be equal. This isrequired so that signaling multiframe integrity is preserved throughoutthe system. In addition, a signal of proper amplitude must be receivedat the HDT 12 to perform any control functions with respect to the ISU100. Likewise, with regard to OFDM transport from the ISUs 100, the ISUs100 must be frequency locked to the HDT 12 such that the multicarrierstransported over the HFC distribution network 11 are orthogonallyaligned. The transport system implements a distributed loop techniquefor implementing this multi-point to point transport utilizingorthogonal frequency division multiplexing as further described below.When the HDT 12 receives the plurality of multi carriers which areorthogonally aligned and which have telephony and control data modulatedthereon with symbols aligned, the MCC modems 82 of the CXMUs 56demodulate the telephony information and control data from the pluralityof multicarriers in their corresponding 6 MHz bandwidth and provide suchtelephony data to the CTSU 54 for delivery to the network interface 62and the control data to the CXMC 80 for control of the telephonytransport.

As one skilled in the art will recognize, the spectrum allocations,frequency assignments, data rates, channel numbers, types of servicesprovided and any other parameters or characteristics of the system whichmay be a choice of design are to be taken as examples only. Theinvention as described in the accompanying claims contemplates suchdesign choices and they therefore fall within the scope of such claims.In addition, many functions may be implemented by software or hardwareand either implementation is contemplated in accordance with the scopeof the claims even though reference may only be made to implementationby one or the other.

First Embodiment of Telephony Transport System

The first embodiment of the telephony transport system in accordancewith the present invention shall be described with particular referenceto FIGS. 13-35 which include block diagrams of MCC modems 82, and HISUmodems and MISU modems shown generally as ISU modem 101 in FIG. 8. Suchmodems implement the upstream and downstream modem transportfunctionality. Following this description is a discussion on the theoryof operation utilizing such modems.

Referring to FIG. 13, the spectrum allocation for one 6 MHz band forupstream and downstream transport of telephony information and controldata utilizing OFDM techniques is shown. The waveform preferably has 240payload channels or DSO+ channels which include 480 carriers or tonesfor accommodating a net data rate of 19.2 Mbps, 24 IOC channelsincluding 46 carriers or tones, and 2 synchronization channels. Eachsynchronization channel includes two carriers or tones and is eachoffset from 24 IOC channels and 240 payload channels by 10 unusedcarriers or tones, utilized as guard tones. The total carriers or tonesis 552. The synchronization tones utilized for synchronization functionsas described further below are located at the ends of the 6 MHz spectrumand the plurality of orthogonal carriers in the 6 MHz band are separatedfrom carriers of adjacent 6 MHz bands by guard bands (516.0 kHz) at eachend of the 6 MHz spectrum. The guard bands are provided at each end ofthe 6 MHz band to allow for filter selectivity at the transmitter andreceivers of the system. The synchronization carriers are offset fromthe telephony data or payload carriers such that if the synchronizationcarrier utilized for synchronization during initialization andactivation is not orthogonal with the other tones or carriers within the6 MHz band, the synchronization signal is prevented from destroying thestructure of the orthogonally aligned waveform. The synchronizationtones are, therefore, outside of the main body of payload carriers ofthe band and interspersed IOC channels, although the synchronizationchannel could be considered a special IOC channel.

To minimize the power requirement of the ISUs, the amount of bandwidththat an ISU processes is minimized. As such, the telephony payloadchannels and IOC channels of the 6 MHz band are interspersed in thetelephony payload channels with an IOC channel located every 10 payloadchannels. With such a distributed technique, wherein subbands of payloadchannels greater than 10 include an IOC channel, the amount of bandwidthan ISU “sees” can be limited such that an IOC channel is available forthe HDT 12 to communicate with the ISU 100. Such subband distributionfor the spectral allocation shown in FIG. 13 is shown in FIG. 16. Thereare 24 subbands in the 6 MHz bandwidth with each subband including 10payload channels with an IOC channel between the 5th and 6th payloadchannels. A benefit of distributing the IOC channels throughout the 6MHz band is protection from narrow band ingress. If ingress destroys anIOC channel, there are other IOC channels available and the HDT 12 canretune an ISU 100 to a different portion of the 6 MHz band, where an IOCchannel that is not corrupted is located. Preferably, the MISU 66 seesapproximately 3 MHz of the 6 MHz bandwidth to receive up to 130 payloadchannels whose bandwidth also includes numerous IOC channels forcommunication from the HDT 12 to the MISU 66. The HISU 68 sees about 100kHz of the 6 MHz bandwidth to receive 11 channels including at least oneIOC channel for communication with the HDT 12.

The primary difference between the downstream and upstream paths are thesupport of downstream synchronization and upstream synchronization. Inthe downstream direction, all ISUs lock to information from the HDT(point to multi-point). The initialization and activation of ISUs arebased on signals supplied in the upstream synchronization channel.During operation, ISUs track the synchronization via the IOC channels.In the upstream, the upstream synchronization process involves thedistributed (multi-point to point) control of amplitude, frequency, andtiming; although frequency control can also be provided utilizing onlythe downstream synchronization channel as described further below. Theprocess of upstream synchronization occurs in one of the two upstreamsynchronization channels, the primary or the secondary synchronizationchannel.

Referring to FIG. 21, the downstream transmission architecture of theMCC modem 82 is shown. Two serial data inputs, approximately 10 Mbpseach, comprise the payload data from the CXMC 56 which is clocked by the8 kHz frame clock input. The IOC control data input from the CXMC 56 isclocked by the IOC clock input, which is preferably a 2.0 kHz clock. Thetelephony payload data and the IOC control data enter through serialports 132 and the data is scrambled as known to one skilled in the artby scrambler 134 to provide randomness in the waveform to be transmittedover the HFC distribution network 11. Without scrambling, very highpeaks in the waveform may occur; however, if the waveform is scrambledthe symbols generated by the MCC modem 82 become sufficiently random andsuch peaks are sufficiently limited.

FIG. 58 details the operation of a typical scrambler, such as 134, FIG.21. Symbol clock 4501 clocks a seed pattern through a linear-feedbackshift register 4510 having nine stages, 4510-0 through 4510-8. With XORgate 4511 positioned as shown, the generator polynomial is binary “100010 000”. The seed initially loaded into register 4510 at input 4502 is“111 001 110”. Two identical translation tables 4520 and 4521 receivetwo-bit inputs from register 4510 at every symbol time. The high- andlow-order bits of table 4520 proceed from the outputs of stages 4510-8and 4510-7, respectively. High-order bit 4523 of table 4521 alsoreceives output 4510-7, but as its high-order bit; stage output 4510-6provides its low-order bit. Logic gate 4530 perform an XOR between thefive-bit output of table 4520 and the upper five bits of a 10-bit DSOword, while gate 4531 does the same for the five-bit output of table4521 and the lower five bits of the same DSO word. Outputs 4505 and 4506carry the two 5-bit scrambled symbols for the DSO word. Each descramblersuch as 176, FIG. 22 or 23, is identical to its corresponding scrambler.It recovers the original bit pattern of each symbol by decoding it withthe same polynomial and seed.

The polynomial and seed for register 4510 of the scramblers anddescramblers selected by known techniques to yield a maximal-lengthpseudo-random sequence. Inversion of the order of the input bits asbetween table 4520 and table 4521 increases the scrambling of the twosymbols of the DSO word. To increase the randomness among differentsequences even more, different scramblers in the system have differentpolynomials and seeds. Randomness could be further increased by usingmore than four different table entries; however, the added complexityoverrode the gain, for this particular embodiment. Only the payloadchannels are scrambled; the IOC channels are not scrambled.

The scrambled signals are applied to a symbol mapping function 136 inFIG. 21. The symbol mapping function 136 takes the input bits and mapsthem into a complex constellation point. For example, if the input bitsare mapped into a symbol for output of a BPSK signal, every bit would bemapped to a single symbol in the constellation as in the mapping diagramfor BPSK of FIG. 15. Such mapping results in in-phase and quadraturevalues (I/Q values) for the data. BPSK is the modulation techniquepreferably used for the upstream and downstream IOC channels and thesynchronization channels. BPSK encoding is preferred for the IOC controldata so as to provide robustness in the system as previously discussed.For QPSK modulation, every two bits would map into one of four complexvalues that represent a constellation point. In the preferredembodiment, 32 QAM is utilized for telephony payload data, wherein everyfive bits of payload data is mapped into one of 32 constellation pointsas shown in FIG. 14. Such mapping also results in I/Q values. As such,one DSO+ signal (10 bits) is represented by two symbols and the twosymbols are transmitted using two carriers. Thus, one DSO+ channel istransported over two carriers or tones of 6 MHz spectrum.

One skilled in the art will recognize that various mapping or encodingtechniques may be utilized with different carriers. For example,telephony channels carrying ISDN may be encoded using QPSK as opposed totelephony channels carrying POTS data being encoded using 32 QAM.Therefore, different telephony channels carrying different services maybe modulated differently to provide for more robust telephony channelsfor those services that require such quality. The architecture inaccordance with the present invention provides the flexibility to encodeand modulate any of the channels differently from the modulationtechnique used for a different channel.

Within the framework of QAM32 modulation, FIG. 17 shows a constellationwhich has improved characteristics. Here, the in-phase and quadraturevalues are shown encoded by three bits each instead of the four shown inFIG. 14; analog values, however, remain in the ranges −5 to +5. Theconstellation of FIG. 17 approaches as closely as possible to an analogyto a Gray code scheme, in which a transition from one row to the nextand from one column to the next result in only a single bit change inthe 5-bit symbol code. (The exceptions are four transitions from thefirst column to the second, and from the fifth to the sixth, which havetwo transitions each. The comer cells have zero transitions betweenthese columns, which do not detract from the advantages of the scheme.)If a symbol is received incorrectly after transmission, the most likelyerror is a slight change in either amplitude or phase. If the bitstrings represented by the symbols have as few bit transitions aspossible for single-value phase and amplitude changes, then a receptionerror will create fewer bit errors on the final digital output. That is,small (symbol) errors in produce small (bit) errors out.

The constellations shown in FIGS. 14 and 17 use all points of a six-cellsquare except the four corners. Hence, they have two axes of symmetry,and appear identical when rotated by 90°, 180°, and 270°. If a phaseerror ever exceeds 45°, an attempted correction may pull the phase to anincorrect orientation. This is called four-fold phase ambiguity.However, deliberately using one and only one of the comer points as avalid symbol provides a key for identifying the correct phase for errorsas great as a full 180°. For example, designating the symbol for “16” asI=010 (+5) and Q=010 (+5) instead of the I=001, Q=010 (+3, +5) in FIG.17 introduces a symbol at this comer point whenever a “16” is sentupstream or downstream. Because only one comer is used, any receivedvalue having both I and Q values ±5 requires phase rotation until I=+5and Q=+5. This assignment also preserves the nearly Gray-code structureof the constellation. Any other symbol assignment which breaks thesymmetry of the constellation would produce the same effect. Even aconstellation retaining only one axis of symmetry would allow twice thephase-correction range of the constellation of FIG. 17. For example,using both the upper left and lower right corners as valid symbolsallows correction of phase errors up to 90°.

Each symbol that gets represented by the I/Q values is mapped into a FFTbin of symbol buffer 138 in FIG. 21. For example, for a DSO+, running at8 kHz frame rate, five bits are mapped into one FFT bin and five bitsinto another bin. Each bin or memory location of the symbol buffer 138represents the payload data and control data in the frequency domain asI/Q values. One set of FFT bins gets mapped into the time domain throughthe inverse FFT 140, as is known to one skilled in the art. The inverseFFT 140 maps the complex I/Q values into time domain samplescorresponding to the number of points in the FFT. Both the payload dataand IOC data are mapped into the buffer 138 and transformed into timedomain samples by the inverse FFT 140. The number of points in theinverse FFT 140 may vary, but in the preferred embodiment the number ofpoints is 256. The output of the inverse FFT 140, for a 256 point FFT,is 256 time domain samples of the waveform.

In conventional practice, buffer 138 clocks symbols into inverse FFT 140at exactly the same rate that inverse FFT 140 clocks out the in-phaseand quadrature values FFT I and FFT Q in FIG. 21. To put the matteranother way, the 256 digital waveform samples from buffer 138 represent360°, or 2n radians, of a QAM 32 waveform having the amplitude and phaseof the 5 bits of its symbol, as determined by mapping unit 136. The FFTI and Q outputs represent 256 samples of a frequency spectrumcorresponding to the same time period. At the receiving end, however,any misalignment at all in the phase synchronization causes FFT 170,FIG. 22, or 180, FIG. 23, to decode a portion of a previous orsubsequent symbol's waveform along with somewhat less than the fullcycle of the desired symbol; this inter-symbol interference can causemisreading the symbol as a different valid symbol, resulting in as manyas five bit errors.

In a presently preferred embodiment, the 256 samples clocked intoinverse FFT 140 represent an extra 45° (n/4 radians) above a completecycle. Another way to think of this is that the symbols are clocked intothe FFT at an effective 9 kHz rate, and clocked out at the nominal 8 kHzsymbol rate. FIG. 52 shows an unmodulated sine wave (that is, one havingI=0, Q=0 in the units used herein). The upper portion shows one cycle,0-360°, at the nominal 8 kHz frame rate. The lower portion shows thesame wave at a 9 kHz rate, so that the amount of time previouslyoccupied by 360° now takes up 405° of phase, from −22.5° to +382.5°.Obviously, there are phase discontinuities between successive cycles ofthe wave. FIG. 53 shows a typical QAM 32 wave modulated at a differentamplitude and a slightly different phase from those of FIG. 52. Thesemight correspond to, say, I=−1, Q=+1 in the scheme used herein. Thesmall portions at the ends of this wave represent un-modulated cycles,as in FIG. 52. The phase of this wave is advanced from the correspondingwave of the lower portion of FIG. 52; it does not cross the zero axis at0° and 180° of its proper cycle. It does, however, include the extra22.5° of excess phase at each end, for 45° extra over an 8 kHz cycle.Again, phase discontinuities exist at the ends of the total 405° phasedegrees of this wave. In fact, this characteristic gives theexcess-phase improvement an advantage over its primary function ofproviding a guard band for the symbol decoder, for reducing inter-symbolinterference.

In FIG. 52, successive cycles of a wave modulated with the same symbol(or with no symbol), produce a continuous waveform with no breaks orother features to distinguish the beginnings and endings of individualcycles. The lower part of this diagram demonstrates that even anun-modulated excess-phase waveform contains discontinuity featuresserving as markers at the ends of each cycle. A repeating string of idlesymbols, or any other symbols, likewise produces these markers. In thefrequency- and phase-acquisition and tracking aspects discussed below,such markers therefore provide definite waveform features forsynchronizing purposes, without having to guarantee the transmission ofany special string of varying characters strictly for synchronization.This saves the overhead of interrupting the payload and/or IOC channelsto provide such a string, and the complexity of storing or divertingpayload information while the sync string is present. It also allowssync to take place at times when, because of the above factors, it wouldnot be feasible otherwise.

At the receiving end, FFT 170 (in an MISU) or 180 (HISU) decodes the 256time slots for one frame time as 405° of a cycle to symbol decoder 174,which matches the cycle to the nearest 5-bit string of bits. Because anyphase difference up to ±22.5° will never conflate the proper wave withthat for another symbol, no inter-symbol interference at all occurswithin this margin of error in phase tracking. This provides a form ofguard band for each symbol. In the upstream direction, units 186, 188,and 190 or 191 provide excess phase in the transmitting MISU and HISUmodems of FIGS. 24 and 25; and the head-end receiving modem of FIG. 26decodes and tracks this phase as described above.

The inverse FFT 140 has separate serial outputs for in-phase andquadrature (I/Q) components, FFT1 and FFTO. Digital to analog converters142 take the in-phase and quadrature components, which is a numericrepresentation of baseband modulated signal and convert it to a discretewaveform. The signal then passes through reconstruction filters 144 toremove harmonic content. This reconstruction is needed to avoid problemsarising from multiple mixing schemes and other filtering problems. Thesignal is summed in a signal conversion transmitter 146 forup-converting the I/Q components utilizing a synthesized waveform thatis digitally tunable with the in-phase and quadrature components formixing to the applicable transmit frequency. For example, if thesynthesizer is at 600 MHz, the output frequency will be at 600 MHz. Thecomponents are summed by the signal conversion transmitter 146 and thewaveform including a plurality of orthogonal carriers is then amplifiedby transmitter amplifier 148 and filtered by transmitter filter 150before being coupled onto the optical fiber by way of telephonytransmitter 14. Such functions are performed under control of generalpurpose processor 149 and other processing circuitry of block 147necessary to perform such modulation. The general purpose processor alsoreceives ISU adjustment parameters from carrier, amplitude, timingrecovery block 222 (FIG. 26) for carrying out distributed loop symbolalignment, frequency locking, amplitude adjustment, and path delayfunctions as described further below.

In conventional practice, the relationship between the frequency of acarrier and the frequency and timing of data symbols modulated onto thatcarrier is arbitrary and unimportant. In the present system, however, ithas been found that even very small frequency drifts between the 8 kHzsymbol or frame clock and the frequencies of the tones upon which theyride can produce significant inter-symbol interference and distortion atthe receiving end. Such drifts tend to destroy the orthogonality of thechannel signals produced by inverse FFT 140 in FIG. 21. The presentsystem also, however, provides a simple, inexpensive way to overcomethis problem. FIG. 51 shows a portion 4200 of the HDT clock/sync logicin CTSU 54, FIG. 3. Timing recovery loop 4210 produces a single masterreference clock output at 10.24 MHz. Although loop 4210 could be afree-running oscillator, it is in fact slaved to the network 10, FIG. 1.With which the entire system communicates. This connection is convenientin eliminating gross or unpredictable differences between the dataspeeds of the network and the system.

Smoothing loop 4220 evens out short-term variations in the signal fromloop 4210. Phase comparator 4221 controls a voltage-controlled crystaloscillator at 40.96 MHz; divider 4223 provides feedback at the properfrequency. Comparator 4221 includes a low-pass integrator which givesphase-lock loop 4220 a bandwidth of about 130 Hz. Divider 4230 reducesthe frequency of VCXO 4222 to 2.56 MHz. A second phase-lock loop 4240has a phase comparator 4241, again with low-pass characteristics,feeding a voltage-controlled oscillator running at 1267.2 MHz; dividerprovides feedback at the proper frequency. Divider 4250 produces thefinal RF clock frequency, 9.9 MHz, at output 4251. The network clock issufficiently accurate over long periods of time, but it is subject tosignificant amounts of short-period jitter. The large amount ofsmoothing provided by loops 4220 and 4240 overcome the intolerance ofanalog RF components for short-term variations. Meanwhile, digitaldivider 4260 divides the master 10.24 MHz clock by a factor of 80 toproduce an 8 kHz symbol or frame clock output 4261. Output 4261 does notrequire the smoothing, because it clocks only digital circuits, whichare relatively insensitive to short-term frequency changes.

RF master clock 4251 proceeds to RF synthesizer 143 in HDT transmittingmodem 82, as shown in FIG. 21, where it directly controls the frequencyof the tunable 500 850 MHz RF carrier for the entire band carrying allof the channels shown in FIGS. 13 and 16. Symbol clock 4261 proceeds tothe frame-clock inputs in FIG. 21, where it controls the symbol timing,and, because it also controls the FFT speed, the frequencies of thechannels in the entire band. Clock lock 4200 thus provides a solid linkwhich inherently preserves the orthogonality of the band signals in amulticarrier system, by deriving the RF carrier clock and the symbol orframe clock from the same source. At the same time, it provides a smallamount of gradual variation for satisfying the demands of the analog RFcomponents.

The overall purpose of locking the two clocks together at the HDT is tolock the carrier clocks and the symbol (frame) clocks throughout thesystem; and the purpose of this in turn is to preserve the orthogonalityof the signals in a multicarrier system which is capable ofbidirectional operation (that is, as a multipoint-to-point-configurationas well as in the usual point-to-multipoint “broadcast” direction).Clock generator 166, FIGS. 22 and 23, of timing generator 107, FIG. 8locks to the frequencies of the incoming signals to provide the clocksused in the remote ISU modules. Therefore, the carrier and frame clocksin each upstream transmitter portion, FIG. 24, of remote modem 108, FIG.8, are also locked to each other, by virtue of being locked to theincoming signal from the HDT.

At the downstream receiving end, either an MISU or an HISU provides forextracting telephony information and control data from the downstreamtransmission in one of the 6 MHz bandwidths. With respect to the MISU 66(FIG. 1), the MISU downstream receiver architecture is shown in FIG. 22.It includes a 100 MHz bandpass filter 152 to reduce the frequency bandof the received 600 to 850 MHz total band broadcast downstream. Thefiltered signal then passes through voltage tuned filters 154 to removeout of band interference and further reduce the bandwidth. The signal isdown converted to baseband frequency via quadrature and in-phase downconverter 158 where the signal is mixed at complex mixers 156 utilizingsynthesizer 157 which is controlled from an output of serial ports 178.The down converted I/Q components are passed through filters 159 andconverted to digital format at analog to digital converters 160. Thetime domain samples of the I/Q components are placed in a sample buffer162 and a set of samples are input to down converter compensation unit164. The compensation unit 164 attempts to mitigate errors such as DCoffsets from the mixers and differential phase delays that occur in thedown conversion.

Carrier, amplitude and timing signaling are extracted from thecompensated signal, by the carrier, amplitude, and timing recovery block166 by extracting control data from the synchronization channels duringinitialization and activation of the ISU and the IOC channels duringtracking as further described below with reference to FIG. 33. Thecompensated signal in parallel form is provided to FFT 170 to beconverted into a vector of frequency domain elements which areessentially the complex constellation points with I/Q componentsoriginally created upstream at the MCC modem 82 for the DSO+ channelswhich the MISU sees. Due to inaccuracies in channel filtering, anequalizer 172 removes dynamic errors that occur during transmission andreception. Equalization in the upstream receiver and the downstreamreceiver architectures shall be explained in further detail below withreference to FIG. 35. From the equalizer 172, the complex constellationpoints are converted to bits by symbol to bit converter 174, descrambledat descrambler 176 which is a mirror element of scrambler 134, and thepayload telephony information and IOC control data are output by theserial ports 178 to the CXSU 102 as shown in FIG. 8. Block 153 includesthe processing capabilities for carrying out the various functions asshown therein.

Referring to FIG. 23, the HISU 68 downstream receiver architecture isshown. The primary difference between the HISU downstream receiverarchitecture (FIG. 23) and the MISU downstream receiver architecture(FIG. 22) is the amount of bandwidth being processed. The front ends ofthe receivers up to the FFT processing are substantially the same,except during the down conversion, the analog to digital converters 160can be operated at a much slower rate. For instance, if the bandwidth ofthe signal being processed is 100 kHz, the sample rate can beapproximately 200 kHz. In an MISU processing a 3 MHz signal, the samplerate is about 6 MHz. Since the HISU is limited to receiving a maximum of10 DSOs, the FFT 180 can be of a smaller size. A 32 point FFT 180 ispreferably used in the HISU and can be implemented more efficiently,compared to a 128 or 256 point FFT utilized in the MISU. Therefore, themajor difference between these architectures is that the HISU receiverarchitecture requires substantially less signal processing capabilitythan the MISU receiver and as such has less power consumption. Thus, toprovide a system wherein power consumption at the remote units isminimized, the smaller band of frequencies seen by the HISU allows forsuch low consumption. One reason the HISU is allowed to see such a smallband of carriers is that the IOC channels are interspersed throughoutthe 6 MHz spectrum.

Referring to FIG. 24, the upstream transmission architecture for theHISU 68 (FIG. 1) is shown. The IOC control data and the telephonypayload data from the CXSU 102 (FIG. 8) is provided to serial ports 182at a much slower rate in the HISU than in the MISU or HDT transmissionarchitectures, because the HISU supports only 10 DSO+ channels. The HISUupstream transmission architecture implements three importantoperations. It adjusts the amplitude of the signal transmitted, thetiming delay (both symbol and path delay) of the signal transmitted, andthe carrier frequency of the signal transmitted. The telephony data andIOC control data enters through the serial ports 182 under control ofclocking signals generated by the clock generator 173 of the HISUdownstream receiver architecture, and is scrambled by scrambler 184 forthe reasons stated above with regard to the MCC downstream transmissionarchitecture. The incoming bits are mapped into symbols, or complexconstellation points, including I/Q components in the frequency domain,by bits to symbol converter 186. The constellation points are thenplaced in symbol buffer 188. Following the buffer 188, an inverse FFT190 is applied to the symbols to create time domain samples; 32 samplescorresponding to the 32 point FFT. A delay buffer 192 is placed on theoutput of the inverse FFT 190 to provide multi-frame alignment at MCCmodem upstream receiver architecture as a function of the upstreamsynchronization process controlled by the HDT 12. The delay buffer 192,therefore, provides a path delay adjustment prior to digital to analogconversion by the digital to analog converters 194 of the in-phase andquadrature components of the output of the inverse FFT 190. Clock delay196 provides a fine tune adjustment for the symbol alignment at therequest of IOC control data output obtained by extracting control datafrom the serial stream of data prior to being scrambled. Afterconversion to analog components by digital to analog converters 194, theanalog components therefrom are reconstructed into a smooth analogwaveform by the reconstruction filters 198. The upstream signal is thendirectly up converted by direct converter 197 to the appropriatetransmit frequency under control of synthesizer block 195. Synthesizerblock 195 is operated under control of commands from an IOC controlchannel which provides carrier frequency adjustment commands thereto asextracted in the HISU downstream receiver architecture. The up convertedsignal is then amplified by transmitter amplifier 200, filtered bytransmitter filter 202 and transmitted upstream to be combined withother signals transmitted by other ISUs 100. The block 181 includesprocessing circuitry for carrying out the functions thereof.

Referring to FIG. 27, the upstream transmitter architecture for the MISU66 is shown and is substantially the same as the upstream transmitterarchitecture of HISU 68. However, the MISU 66 handles more channels andcannot perform the operation on a single processor as can the HISU 68.Therefore, both a processor of block 181 providing the functions ofblock 181 including the inverse FFT 190 and a general purpose processor206 to support the architecture are needed to handle the increasedchannel capacity.

Referring to FIG. 26, the MCC upstream receiver architecture of eachCXMU 56 at the HDT 12 is shown. A 5 to 40 MHz band pass filter 208filters the upstream signal which is then subjected to a direct downconversion to baseband by mixer and synthesizer circuitry 211. Theoutputs of the down conversion is applied to anti-alias filters 210 forconditioning thereof and the output signal is converted to digitalformat by analog to digital converters 212 to provide a time domainsampling of the in-phase and quadrature components of the signal tonarrow band ingress filter and FFT 112. The narrow band ingress filterand FFT 112, as described below, provides protection against narrow bandinterference that may affect the upstream transmission.

The ingress filter and FFT 112 protects ten channels at a time,therefore, if ingress affects one of the available 240 DSO+s in the 6MHz spectrum received by MCC modem 82, a maximum of ten channels will bedestroyed from the ingress. The ingress filter and FFT 112 includes apolyphase structure, as will be recognized by one skilled in the art asa common filter technique. It will be further recognized by one skilledin the art that the number of channels protected by the polyphase filtercan be varied. The output of the ingress filter and FFT 112 is coupledto an equalizer 214 which provides correction for inaccuracies thatoccur in the channel, such as those due to noise from referenceoscillators or synthesizers. The output symbols of the equalizer 214,are applied to a symbols to bits converter 216 where the symbols aremapped into bits. The bits are provided to descramblers 218, which are amirror of the scramblers of the ISUs 100 and the output of thedescramblers are provided to serial ports 220. The output of the serialports is broken into two payload streams and one IOC control data streamjust as is provided to the MCC downstream transmitter architecture inthe downstream direction. Block 217 includes the necessary processingcircuitry for carrying out the functions therein.

In order to detect the downstream information, the amplitude, frequency,and timing of the arriving signal must be acquired using the downstreamsynchronization process. Since the downstream signal constitutes a pointto multi-point node topology, the OFDM waveform arrives via a singlepath in an inherently synchronous manner, in contrast to the upstreamsignal. Acquisition of the waveform parameters is initially performed onthe downstream synchronization channels in the downstreamsynchronization bands located at the ends of the 6 MHz spectrum. Thesesynchronization bands include a single synchronization carrier or tonewhich is BPSK modulated by a 2 kHz framing clock. This tone is used toderive initial amplitude, frequency, and timing at the ISU. Thesynchronization carrier may be located in the center of the receive bandand could be considered a special case of an IOC. After the signal isreceived and the receiver architecture is tuned to a typical IOCchannel, the same circuitry is used to track the synchronizationparameters using the IOC channel. The process used to acquire thenecessary signal parameters utilizes carrier, amplitude and timingrecovery block 166 of the ISU receiver architecture (FIG. 22), which isshown in more detail in block diagram form in FIG. 33. The carrier,amplitude and timing recovery block 166 includes a Costas loop 330 whichis used to acquire the frequency lock for the received waveform. Afterthe signal is received from the compensation unit 164 (FIG. 22), asample and hold 334 and analog to digital conversion 332 is applied tothe signal with the resulting samples from the converters 332 applied tothe Costas loop 330. The sampling is performed under control of voltagecontrolled oscillator 340 as divided by divider 333 which divides by thenumber of points of the FFT utilized in the receiver architecture, M.The mixers 331 of the Costas loop 330 are fed by the arriving signal andthe feedback path, and serve as the loop phase detectors. The output ofthe mixers 331 are filtered and decimated to reduce the processingrequirements of subsequent hardware. Given that the received signal isband-limited, less samples are required to represent the synchronizationsignal. If orthogonality is not preserved in the receiver, the filterwill eliminate undesired signal components from the recovery process.Under conditions of orthogonality, the LPF 337 will completely removeeffects from adjacent OFDM carriers. When carrier frequency lock isachieved, the process will reveal the desired BPSK waveform in thein-phase arm of the loop. The output of the decimators are fed throughanother mixer, then processed through the loop filter with filterfunction H(s) and numerically controlled oscillator (NCO), completingthe feedback path to correct for frequency error. When the error is at a“small” level, the loop is locked. In order to achieve fast acquisitionand minimal jitter during tracking, it will be necessary to employ dualloop bandwidths. System operation will require that frequency lock isachieved and maintained within about ±4% of the OFDM channel spacing(360 Hz). The amplitude of the signal is measured at the output of thefrequency recovery loop at BPSK power detector 336. The total signalpower will be measured, and can be used to adjust a numericallycontrollable analog gain circuit (not shown). The gain circuit isintended to normalize the signal so that the analog to digitalconverters are used in an optimal operating region.

Timing recovery is performed using an early-late gate type algorithm ofearly-late gate phase detector 338 to derive timing error, and byadjusting the sample clock or oscillator 340 in response to the errorsignal. The early-late gate detector results in an advance/retardcommand during an update interval. This command will be applied to thesample clock or oscillator 340 through filter 341. This loop is held offuntil frequency lock and amplitude lock have been achieved. When thetiming loop is locked, it generates a lock indicator signal. The sameclocks are also used for the upstream transmission. The carrier, timingand amplitude recovery block 166 provides a reference for the clockgenerator 168. The clock generator 168 provides all of the clocks neededby the MISU, for example, the 8 kHz frame clock and the sample clock.

Carrier, amplitude, and timing recovery block 222 of the MCC modemupstream receiver architecture (FIG. 26), is shown by thesynchronization loop diagram of FIG. 34. It performs detection forupstream synchronization on signals on the upstream synchronizationchannel. For initialization and activation of an ISU, upstreamsynchronization is performed by the HDT commanding one of the ISUs viathe downstream IOC control channels to send a reference signal upstreamon a synchronization channel. The carrier, amplitude, and timingrecovery block 222 measures the parameters of data from the ISU 100 thatresponds on the synchronization channel and estimates the frequencyerror, the amplitude error, and the timing error compared to referencesat the HDT 12. The output of the carrier, amplitude, and timing recoveryblock 222 is turned into adjustment commands by the HDT 12 and sent tothe ISU being initialized and activated in the downstream direction onan IOC control channel by the MCC downstream transmitter architecture.

The purpose of the upstream synchronization process is to initialize andactivate ISUs such that the waveform from distinct ISUs combine to aunified waveform at the HDT 12. The parameters that are estimated at theHDT 12 by carrier, amplitude, and timing recovery block 222 and adjustedby the ISUs are amplitude, timing, and frequency. The amplitude of anISUs signal is normalized so that DSO+s are apportioned an equal amountof power, and achieves a desired signal to noise ratio at the HDT 12. Inaddition, adjacent ISUs must be received at the correct relative levelor else weaker DSO+ channels will be adversely impacted by the transientbehavior of the stronger DSO+ channels. If a payload channel istransmitted adjacent to another payload channel with sufficientfrequency error, orthogonality in the OFDM waveform deteriorates anderror rate performance is compromised. Therefore, the frequency of theISU must be adjusted to close tolerances. Timing of the recovered signalalso impacts orthogonality. A symbol which is not aligned in time withadjacent symbols can produce transitions within the part of the symbolthat is subjected to the FFT process. If the transitions of all symbolsdon't fall within the guard interval at the HDT, approximately ±16 tones(8 DSO+s) relative to the non-orthogonal channel will be unrecoverable.

During upstream synchronization, the ISUs will be commanded to send asignal, for example a square wave signal, to establish amplitude andfrequency accuracy and to align symbols. The pattern signal may be anysignal which allows for detection of the parameters by carrier,amplitude and timing recovery block 222 and such signal may be differentfor detecting different parameters. For example, the signal may be acontinuous sinusoid for amplitude and frequency detection and correctionand a square wave for symbol timing. The carrier, amplitude and timingrecovery block 222 estimates the three distributed loop parameters. Inall three loops, the resulting error signal will be converted to acommand by the CXMC 80 and sent via the MCC modem 82 over an IOC channeland the CXSU will receive the command and control the adjustment made bythe ISU.

As shown in FIG. 34, the upstream synchronization from the ISU issampled and held 434 and analog to digital converted 432 under controlof voltage controlled oscillator 440. Voltage controlled oscillator is alocal reference oscillator which is divided by M, the points of the FFTin the receiver architecture, for control of sample and hold 434 andanalog to digital converter 432 and divided by k to apply an 8 kHzsignal to phase detector 438.

Frequency error may be estimated utilizing the Costas loop 430. TheCostas loop 430 attempts to establish phase lock with the locallygenerated frequency reference. After some period of time, loopadaptation will be disabled and phase difference with respect to thetime will be used to estimate the frequency error. The frequency erroris generated by filter function H(s) 444 and provided to the CXMC 80 forprocessing to send a frequency adjustment command to the ISU via an IOCcontrol channel. The frequency error is also applied to the numericallycontrolled oscillator (NCO) to complete the frequency loop to correctfor frequency error.

The amplitude error is computed based on the magnitude of the carrierduring the upstream synchronization by detecting the carrier amplitudeof the in-phase arm of the Costas loop 430 by power detector 436. Theamplitude is compared with a desired reference value at referencecomparator 443 and the error will be sent to the CXMC 80 for processingto send an amplitude adjustment command to the ISU via an IOC controlchannel.

When the local reference in the HDT has achieved phase lock, the BPSKsignal on the synchronization channel arriving from the ISU is availablefor processing. The square wave is obtained on the in-phase arm of theCostas loop 430 and applied to early-late gate phase detector 438 forcomparison to the locally generated 8 kHz signal from divider 435. Thephase detector 438 generates a phase or symbol timing error applied toloop filter 441 and output via line 439. The phase or symbol timingerror is then provided to the CXMC 80 for processing to send a symboltiming adjustment command to the ISU via an IOC control channel.

The mechanisms in the ISU which adjust the parameters for upstreamsynchronization include implementing an amplitude change with a scalarmultiplication of the time domain waveform as it is being collected fromthe digital processing algorithm, such as inverse FFT 190, by thedigital to analog converters 194 (FIG. 24). Similarly, a complex mixingsignal could be created and implemented as a complex multiply applied tothe input to the digital to analog converters 194. Frequency accuracy ofboth the downstream sample clock and upstream sample clock, in the ISU,is established by phase locking an oscillator to the downstreamsynchronization and IOC information. Upstream transmission frequency isadjusted, for example, at synthesizer block 195 as commanded by the HDT12.

Symbol timing corrections are implemented as a delay function. Symboltiming alignment in the ISU upstream direction is therefore establishedas a delay in the sample timing accomplished by either blanking a sampleinterval (two of the same samples to go out simultaneously) or byputting in an extra clock edge (one sample is clocked out and lost) viaclock delay 196 (FIG. 24). In this manner, a delay function can becontrolled without data storage overhead beyond that already required.

After the ISU is initialized and activated into the system, ready fortransmission, the ISU will maintain required upstream synchronizationsystem parameters using the carrier, amplitude, frequency recovery block222. An unused but initialized and activated ISU will be commanded totransmit on an IOC and the block 222 will estimate the parameterstherefrom as explained above.

In both the upstream transmitter architectures for the MISU 66 (FIG. 24)and the HISU 68 (FIG. 25), frequency offset or correction to achieveorthogonality of the carriers at HDT 12 can be determined on the ISU asopposed to the frequency offset being determined at the HDT duringsynchronization by carrier, amplitude and timing recovery block 222(FIG. 26) and then frequency offset adjustment commands beingtransmitted to the ISU for adjustment of carrier frequency via thesynthesizer blocks 195 and 199 of the HISU 68 and MISU 66, respectively.Thus, frequency error would no longer be detected by carrier, amplitudeand timing recovery block 222 as described above. Rather, in such adirect ISU implementation, the ISU, whether an HISU 68 or MISU 66,estimates a frequency error digitally from the downstream signal and acorrection is applied to the upstream data being transmitted.

The HDT 12 derives all transmit and receive frequencies from the samefundamental oscillator. Therefore, all mixing signals are frequencylocked in the HDT. Similarly, the ISU, whether an HISU 68 or MISU 66,derives all transmit and receive frequencies from the same fundamentaloscillator; therefore, all the mixing signals on the ISU are alsofrequency locked. There is, however, a frequency offset present in theISU oscillators relative to the HDT oscillators. The amount of frequencyerror (viewed from the ISU) will be a fixed percentage of the mixingfrequency. For example, if the ISU oscillator is 1OPPM off in frequencyfrom the HDT oscillators, and the downstream ISU receiver mix frequencywas 100 MHz and the ISU upstream transmit mixing frequency were 10 MHz,the ISU would have to correct for 1 kHz on the downstream receiver andcreate a signal with a 100 Hz offset on the upstream transmitter. Assuch, with the ISU direct implementation, the frequency offset isestimated from the downstream signal.

The estimation is performed with digital circuitry performing numericcalculations (that is, a processor). Samples of the synchronizationchannel or IOC channel are collected in hardware during operation of thesystem. A tracking loop drives a digital numeric oscillator which isdigitally mixed against the received signal. This process derives asignal internally that is essentially locked to the HDT. The internalnumerical mix accounts for the frequency offset. During the process oflocking to the downstream signal in the ISU, the estimate of frequencyerror is derived and with the downstream frequency being known, afractional frequency error can be computed. Based on the knowledge ofthe mixing frequency at the HDT that will be used to down convert theupstream receive signal, an offset to the ISU transmit frequency iscomputed. This frequency offset is digitally applied to the ISUtransmitted signal prior to converting the signal to the analog domain,such as by converters 194 of FIG. 24. Therefore, the frequencycorrection can be performed directly on the ISU.

Referring to FIGS. 31 and 32, the narrow band ingress filter and FFT 112of the MCC upstream receiver architecture, including a polyphase filterstructure, will be described in further detail. Generally, the polyphasefilter structure includes polyphase filters 122 and 124 and providesprotection against ingress. The 6 MHz band of upstream OFDM carriersfrom the ISUs 100 is broken into subbands through the polyphase filterswhich provide filtering for small groups of carriers or tones, and if aningress affects carriers within a group of carriers, only that group ofcarriers is affected and the other groups of carriers are protected bysuch filtering characteristics.

The ingress filter structure has two parallel banks 122, 124 ofpolyphase filters. One bank has approximately 17 differentnon-overlapping bands with channel spaces between the bands. A magnituderesponse of a single polyphase filter bank is shown in FIG. 29. Thesecond bank is offset from the first bank by an amount so that thechannels that are not filtered by the first bank are filtered by thesecond bank. Therefore, as shown in the close-up magnitude response of asingle polyphase filter bank in FIG. 30, one band of channels filteredmay include those in frequency bins 38-68 with the center carrierscorresponding to bins 45-61 being passed by the filter. The overlappingfilter provides for filtering carriers in the spaces between the bandsand the carriers not passed by the other filter bank. For example, theoverlapping filter may pass 28-44. The two channel banks are offset by16 frequency bins so that the combination of the two filter banksreceives every one of the 544 channels.

Referring to FIG. 31, the ingress filter structure receives the sampledwaveform x(k) from the analog to digital converters 212 and then complexmixers 118 and 120 provide the stagger for application to the polyphasefilters 122, 124. The mixer 118 uses a constant value and the mixer 120uses a value to achieve such offset. The outputs of each mixer entersone of the polyphase filters 122, 124. The output of each polyphasefilter bank comprises 18 bands, each of which contain 16 usable FFT binsor each band supports sixteen carriers at the 8 kHz rate, or 8 DSO+'s.One band is not utilized.

Each band output of the polyphase filters 122, 124 has 36 samples per 8kHz frame including 4 guard samples and enters a FFT block 126, 128. Thefirst operation performed by the FFT blocks 126, 128 is to remove thefour guard samples, thereby leaving 32 time domain points. The output ofeach FFT in the blocks is 32 frequency bins, 16 of which are used withthe other bins providing filtering. The output of the FFTs are staggeredto provide overlap. As seen in FIG. 31, carriers 0-15 are output by FFT#1 of the top bank, carriers 16-31 are output by FFT #1 of the bottombank, carriers 32-48 are output by FFT #2 of the top bank and so on.

The polyphase filters 122, 124 are each standard polyphase filterconstruction as is known to one skilled in the art and each is shown bythe structure of FIG. 32. The input signal is sampled at a 5.184mega-sample per second rate, or 648 samples per frame. The input is thendecimated by a factor of 18 (1 of 18 samples are kept) to give aneffective sample rate of 288 kHz. This signal is subjected to the finiteimpulse response (FIR) filters, labeled H0,0(Z) through H_(o),₁₆(Z),which include a number of taps, preferably 5 taps per filter. As oneskilled in the art will recognize the number of taps can vary and is notintended to limit the scope of the invention. The outputs from thefilters enter an 18 point inverse FFT 130. The output of the transformis 36 samples for an 8 kHz frame including 4 guard samples and isprovided to FFT blocks 126 and 128 for processing as described above.The FFT tones are preferably spaced at 9 kHz, and the information rateis 8 kilo-symbols per second with four guard samples per symbolallotted. The 17 bands from each polyphase filter are applied to the FFTblocks 126, 128 for processing and output of the 544 carriers asindicated above. One band, the 18^(th) band, as indicated above, is notused.

The equalizer 214 (FIG. 26) and 172 (FIG. 22), in both upstream anddownstream receiver architectures, is supplied to account for changes ingroup delay across the cable plant. The equalizer tracks out phase andgain or amplitude variations due to environmental changes and cantherefore adapt slowly while maintaining sufficiently accurate tracking.The coefficients 360 of the equalizer 172,214, for which the internalequalizer operation is generally shown in FIG. 35, represent the inverseof the channel frequency response to the resolution of the FFT 112, 170.The downstream coefficients will be highly correlated since everychannel will progress through the same signal path as opposed to theupstream coefficients which may be uncorrelated due to the variantchannels that individual DSO+s may encounter in the multi-point to pointtopology. While the channel characteristics are diverse, the equalizerwill operate the same for either upstream or downstream receivers. Thedownstream equalizer will track on only the IOC channels, thus reducingthe computational requirements at the ISUs and removing the requirementfor a preamble in the payload channels, as described further below,since the IOC channels are always transmitted. The upstream, however,will require equalization on a per DSO+ and IOC channel basis.

The algorithm used to update the equalizer coefficients contains severallocal minima when operating on a 32 QAM constellation and suffers from afour-fold phase ambiguity. Furthermore, each DSO+ in the upstream canemanate from a separate ISU, and can therefore have an independent phaseshift. To mitigate this problem, each communication onset will berequired to post a fixed symbol preamble prior to data transmission.Note that the IOC channels are excluded from this requirement since theyare not equalized and that the preamble cannot be scrambled. It is knownthat at the time of transmission, the HDT 12 will still have accuratefrequency lock and symbol timing as established during initializationand activation of the ISU and will maintain synchronization on thecontinuously available downstream IOC channel.

The introduction of the preamble requires that the equalizer haveknowledge of its process state. Three states are introduced whichinclude: search, acquisition, and tracking mode. Search mode is based onthe amount of power present on a channel. Transmitter algorithms willplace a zero value in unused FFT bins, resulting in no power beingtransmitted on that particular frequency. At the receiver, the equalizerwill determine that it is in search mode based on the absence of powerin the FFT bin.

When transmission begins for an initialized and activated ISU, theequalizer detects the presence of signal and enters the acquisitionmode. The length of the preamble may be about 15 symbols. The equalizerwill vary the equalization process based on the preamble. The initialphase and amplitude correction will be large but subsequent updates ofthe coefficients will be less significant. In order to differentiate thetraining pattern from any other data sequence, when the HDT informs anISU to connect a new payload channel, the ISU transmits 16 consecutivesymbols having I=0 and Q=0, which is not a valid data symbol in theconstellations of FIG. 14 or 17. The ISU then transmits 8 valid datasymbols, allowing the equalizer for that channel to set its coefficientproperly to adjust for amplitude and phase of the incoming signal.

After acquisition, the equalizer will enter a tracking mode with theupdate rate being reduced to a minimal level. The tracking mode willcontinue until a loss of power is detected on the channel for a periodof time. The channel is then in the unused but initialized and activatedstate. The equalizer will not train or track when the receiver is beingtuned and the coefficients will not be updated. The coefficients may beaccessed and used such as by signal to noise detector 305 (FIG. 26) forchannel monitoring as discussed further below.

For the equalization process, the I/Q components are loaded into abuffer at the output of the FFT, such as FFT 112, 180. As will beapparent to one skilled in the art, the following description of theequalizer structure is with regard to the upstream receiver equalizer214 but is equally applicable to the downstream receiver equalizer 172.The equalizer 214 extracts time domain samples from the buffer andprocesses one complex sample at a time. The processed information isthen output therefrom. FIG. 35 shows the basic structure of theequalizer algorithm less the state control algorithm which should beapparent to one skilled in the art. The primary equalization pathperforms a complex multiply at multiplier 370 with the value from theselected FFT bin. The output is then quantized at symbol quantize block366 to the nearest symbol value from a storage table. The quantizedvalue (hard decision) is passed out to be decoded into bits by symbolsto bits converter 216. The remainder of the circuitry is used to updatethe equalizer coefficients. An error is calculated between the quantizedsymbol value and the equalized sample at summer 364. This complex erroris multiplied with the received sample at multiplier 363 and the resultis scaled by the adaptation coefficient by multiplier 362 to form anupdate value. The update value is summed at summer 368 with the originalcoefficient to result in a new coefficient value.

Operation of First Embodiment

In the preferred embodiment, the 6 MHz frequency band for each MCC modem82 of HDT 12 is allocated as shown in FIG. 13. Although the MCC modem 82transmits and receives the entire 6 MHz band, the ISU modems 100 (FIG.8) are optimized for the specific application for which they aredesigned and may terminate/generate fewer than the total number ofcarriers or tones allocated in the 6 MHz band. The upstream anddownstream band allocations are preferably symmetric. The upstream 6 MHzbands from the MCC modems 82 lie in the 5-40 MHz spectrum and thedownstream 6 MHz bands lie in the 725-760 MHz spectrum. One skilled inthe art will recognize that if different transmission media are utilizedfor upstream and downstream transmission, the frequencies fortransmission may be the same or overlap but still be non-interfering.

There are three regions in each 6 MHz frequency band to support specificoperations, such as transport of telephony payload data, transport ofISU system operations and control data (IOC control data), and upstreamand downstream synchronization. Each carrier or tone in the OFDMfrequency band consists of a sinusoid which is modulated in amplitudeand phase to form a complex constellation point as previously described.The fundamental symbol rate of the OFDM waveform is 8 kHz, and there area total of 552 tones in the 6 MHz band. The following Table 2 summarizesthe preferable modulation type and bandwidth allocation for the varioustone classifications.

TABLE 2 Band Number of Tones or Allocation Carriers Modulation CapacityBandwidth Synch Band 24 tones (2 synch tones BPSK n/a 216 kHz at eachend and 10 guard tones at each end) Payload Data 480 (240 DS0+ 32 QAM19.2 MBPS 4.32 mhZ channels) IOC 48 (2 every 20 data BPSK 384 kbps 432kHz channels or 24 IOC channels) Intra-band Remainder on each end n/an/a 1.032 MHz Guard (516 kHz at each end) Composite 552 n/a n/a 6.0 MHzSignal

Guard bands are provided at each end of the spectrum to allow forselectivity filtering after transmission and prior to reception. A totalof 240 telephony data channels are included throughout the band, whichaccommodates a net data rate of 19.2 Mbps. This capacity was designed toaccount for additive ingress, thereby retaining enough support toachieve concentration of users to the central office. The IOC channelsare interspersed throughout the band to provide redundancy andcommunication support to narrowband receivers located in the HISUs. TheIOC data rate is 16 kbps (two BPSK tones at the symbol rate of 8 kHzframes per second). Effectively, an IOC is provided for every 10 payloaddata channels. An ISU, such as an HISU, that can only see a single IOCchannel would be forced to retune if the IOC channel is corrupted.However, an ISU which can see multiple IOC channels can select analternate IOC channel in the event that the primary choice is corrupt,such as for an MISU.

The synchronization channels are duplicated at the ends of the band forredundancy, and are offset from the main body of usable carriers toguarantee that the synchronization channels do not interfere with theother used channels. The synchronization channels were previouslydescribed and will be further described below. The synchronizationchannels are operated at a lower power level than the telephony payloadchannels to also reduce the effect of any interference to such channels.This power reduction also allows for a smaller guard band to be usedbetween the synchronization channels and the payload telephony channels.One synchronization or redundant synchronization channels may also beimplemented within the telephony channels as opposed to being offsettherefrom. In order to keep them from interfering with the telephonychannels, the synchronization channels may be implemented using a lowersymbol rate. For example, if the telephony channels are implemented atan 8 kHz symbol rate, the synchronization channels could be implementedat a 2 kHz symbol rate and also may be at a lower power level.

The ISUs 100 are designed to receive a subband, as shown in FIG. 16, ofthe total aggregate 6 MHz spectrum. As an example, the HISU 68 willpreferably detect only 22 of the available 552 channels. Thisimplementation is primarily a cost/power savings technique. By reducingthe number of channels being received, the sample rate and associatedprocessing requirements are dramatically reduced and can be achievablewith common conversion parts on the market today.

A given HISU 68 is limited to receiving a maximum of 10 DSOs out of thepayload data channels in the HISU receiver's frequency view. Theremaining channels will be used as a guard interval. Furthermore, inorder to reduce the power/cost requirements, synthesizing frequencysteps will be limited to 198 kHz. An IOC channel is provided for asshown in FIG. 16 so that every HISU 68 will always see an IOC channelfor control of the HISU 68 via HDT 12. The MISU 66 is designed toreceive 13 subbands, as shown in FIG. 16, or 130 of the 240 availableDSOs. Again, the tuning steps will be limited to 198 kHz to realize anefficient synthesizer implementation. These are preferred values for theHISU 68 and MISU 66, and it will be noted by one skilled in the art thatmany of the values specified herein can be varied without changing thescope or spirit of the invention as defined by the accompanying claims.

As known to one skilled in the art, there may be need to supportoperation over channels in a bandwidth of less than 6 MHz. Withappropriate software and hardware modifications of the system, suchreconfiguration is possible as would be apparent to one skilled in theart. For example, for a 2 MHz system, in the downstream, the HDT 12would generate the channels over a subset of the total band. The HISUsare inherently narrowband and would be able to tune into the 2 MHz band.The MISUs supporting 130 channels, would receive signals beyond the 2MHz band. They would require reduction in filter selectivity by way of ahardware modification. An eighty (80) channel MISU would be able tooperate with the constraints of the 2 MHz. system. In the upstream, theHISUs would generate signals within the 2 MHz band and the MISUstransmit section would restrict the information generated to thenarrower band. At the HDT, the ingress filtering would providesufficient selectivity against out of band signal energy. The narrowbandsystem would require synchronization bands at the edges of the 2 MHzband.

As previously described, acquisition of signal parameters forinitializing the system for detection of the downstream information isperformed using the downstream synchronization channels. The ISUs usethe carrier, amplitude, timing recovery block 166 to establish thedownstream synchronization of frequency, amplitude and timing for suchdetection of downstream information. The downstream signal constitutes apoint to multi-point topology and the OFDM waveform arrives at the ISUsvia a single path in an inherently synchronous manner. In the upstreamdirection, each ISU 100 must be initialized and activated through aprocess of upstream synchronization before an HDT 12 can enable the ISU100 for transmission. The process of upstream synchronization for theISUs is utilized so that the waveform from distinct ISUs combine to aunified waveform at the HDT. The upstream synchronization process,portions of which were previously described, involves various steps.They include: ISU transmission level adjustment, upstream multicarriersymbol alignment, carrier frequency adjustment, and round trip pathdelay adjustment. Such synchronization is performed after acquisition ofa 6 MHz band of operation.

Generally, with respect to level adjustment, the HDT 12 calibrates themeasured signal strength of the upstream transmission received from anISU 100 and adjusts the ISU 100 transmit level so that all ISUs arewithin acceptable threshold. Level adjustment is performed prior tosymbol alignment and path delay adjustment to maximize the accuracy ofthese measurements.

Generally, symbol alignment is a necessary requirement for themulticarrier modulation approach implemented by the MCC modems 82 andthe ISU modems 101. In the downstream direction of transmission, allinformation received at the ISU 100 is generated by a single CXMU 56, sothe symbols modulated on each multicarrier are automatically phasealigned. However, upstream symbol alignment at the MCC modem 82 receiverarchitecture varies due to the multi-point to point nature of the HFCdistribution network 11 and the unequal delay paths of the ISUs 100. Inorder to have maximum receiver efficiency, all upstream symbols must bealigned within a narrow phase margin. This is done by providing anadjustable delay path parameter in each ISU 100 such that the symbolperiods of all channels received upstream from the different ISUs arealigned at the point they reach the HDT 12.

Generally, round trip path delay adjustment is performed such that theround trip delay from the HDT network interface 62 to all ISUs 100 andback to the network interface 62 from all the ISUs 100 in a system mustbe equal. This is required so that signaling multiframe integrity ispreserved throughout the system. All round trip processing for thetelephony transport section has a predictable delay with the exceptionof the physical delay associated with signal propagation across the HFCdistribution network 11 itself ISUs 100 located at close physicaldistance from the HDT 12 will have less round trip delay than ISUslocated at the maximum distance from the HDT 12. Path delay adjustmentis implemented to force the transport system of all ISUs to have equalround trip propagation delay. This also maintains DS1 multiframealignment for DS1 channels transported through the system, maintainingin-band channel signaling or robbed-bit signaling with the samealignment for voice services associated with the same DS1.

Generally, carrier frequency adjustment must be performed such that thespacing between carrier frequencies is such as to maintain orthogonalityof the carriers. If the multicarriers are not received at the MCC modem82 in orthogonal alignment, interference between the multicarriers mayoccur. Such carrier frequency adjustment can be performed in a mannerlike that of symbol timing or amplitude adjustment or may be implementedon the ISU as described previously above.

In the initialization process, when the ISU has just been powered up2710, the ISU 100 has no knowledge of which downstream 6 MHz frequencyband it should be receiving in. This results in the need for theacquisition of 6 MHz band in the initialization process. Until an ISU100 has successfully acquired a 6 MHz band for operation, it implementsa “scanning” approach to locate its downstream frequency band.Generally, a local processor of the CXSU controller 102 of ISU 100starts with a default 6 MHz receive frequency band somewhere in therange from 625 to 850 MHz. The ISU 100 waits for a period of time, forexample 100 milliseconds, in each 6 MHz band to look for a valid 6 MHzacquisition command which matches a unique identification number for theISU 100 after obtaining a valid synchronization signal; which uniqueidentifier may take the form of or be based on a serial number of theISU equipment. If a valid 6 MHz acquisition command or validsynchronization command is not found in that 6 MHz band, the CXSUcontroller 102 looks at the next 6 MHz band and the process is repeated.In this manner, as explained further below, the HDT 12 can tell the ISU100 which 6 MHz band it should use for frequency reception and laterwhich band for frequency transmission upstream.

The process of initialization and activation of ISUs, as generallydescribed above, and tracking or follow-up synchronization is furtherdescribed below. This description is written using an MISU 66 inconjunction with a CXSU controller 102 but is equally applicable to anyISU 100 implemented with an equivalent controller logic. The coax mastercard logic (CXMC) 80 is instructed by the shelf controller unit (SCNU)58 to initialize and activate a particular ISU 100. The SCNU uses an ISUdesignation number to address the ISU 100. The CXMC 80 correlates theISU designation number with an equipment serial number, or uniqueidentifier, for the equipment. No two ISU equipments shipped from thefactory possess the same unique identifier. If the ISU 100 has neverbefore been initialized and activated in the current system database,the CXMC 80 chooses a personal identification number (PIN) code for theISU 100 being initialized and activated. This PIN code is then stored inthe CXMC 80 and effectively represents the “address” for allcommunications with that ISU 100 which will follow. The CXMC 80maintains a lookup table between each ISU designation number, the uniqueidentifier for the ISU equipment, and the PIN code. Each ISU 100associated with the CXMU 56 has a unique PIN address code assignment.One PIN address code will be reserved for a broadcast feature to allISUs, which allows for the HDT to send messages to all initialized andactivated ISUs 100.

The CXMC 80 sends an initialization and activation enabling message tothe MCC modem 82 which notifies the MCC modem 82 that the process isbeginning and the associated detection functionality in the MCC modem 82should be enabled. Such functionality is performed at least in part bycarrier, amplitude, timing recovery block 222 as shown in the MCCupstream receiver architecture of FIG. 26 and as previously discussed.

The CXMC 80 transmits an identification message by the MCC modem 82 overall IOC channels of the 6 MHz band in which it transmits. The messageincludes a PIN address code to be assigned to the ISU being initializedand activated, a command indicating that ISU initialization andactivation should be enabled at the ISU 100, the unique identifier forthe ISU equipment, such as the equipment serial number, and cyclicalredundancy checksum (CRC). The messages are sent periodically for acertain period of time, TscAN, which is shown as 6.16 seconds in FIG. 20and which is also shown in FIG. 19. This period of time is the maximumtime which an ISU can scan all downstream 6 MHz bands, synchronize, andlisten for a valid identification message. The periodic rate, forexample 50 ms, affects how quickly the ISU learns its identity. The CXMC80 will never attempt to synchronize more than one ISU at a time, butwill attempt to identify several ISUs during burst identification asdescribed further below. A software timeout is implemented if an ISUdoes not respond after some maximum time limit is exceeded. This timeoutmust be beyond the maximum time limit required for an ISU to obtainsynchronization functions.

During periodic transmission by CXMC 80, the ISU implements the scanningapproach to locate its downstream frequency band. The local processor ofthe CXSU starts with a default 6 MHz receive frequency band somewhere inthe range from 625 to 850 MHz. The ISU 100 selects the primarysynchronization channel of the 6 MHz band and then tests for loss ofsynchronization after a period of time. If loss of synchronization isstill present, the secondary synchronization channel is selected andtested for loss of synchronization after a period of time. If loss ofsynchronization is still present, then the ISU restarts selection of thesynchronization channels on the next 6 MHz band which may be 1 MHz awaybut still 6 MHz in width. When loss of synchronization is not present ona synchronization channel then the ISU selects the first subbandincluding the IOC and listens for a correct identification message. If acorrect identification message is found which matches its uniqueidentifier then the PIN address code is latched into an appropriateregister. If a correct identification message is not found in the firstsubband on that IOC then a middle subband and IOC is selected, such asthe 11th subband, and the ISU again listens for the correctidentification message. If the message again is not correctly detected,then the ISU restarts on another 6 MHz band. The ISU listens for thecorrect identification message in a subband for a period of time equalto at least two times the CXMU transmission time, for example 100 mswhen transmission time is 50 ms as described above. The identificationcommand is a unique command in the ISU 100, as the ISU 100 will notrequire a PIN address code match to respond to such commands, but only avalid unique identifier and CRC match. If an uninitialized andun-activated ISU 100 receives an identification command from the CXMC 80via the MCC modem 82 on an IOC channel, data which matches the uniqueidentifier and a valid CRC, the CXSU 102 of the ISU 100 will store thePIN address code transmitted with the command and unique identifier.From this point on, the ISU 100 will only respond to commands whichaddress it by its correct PIN address code, or a broadcast address code;unless, of course, the ISU is re-activated again and given a new PINaddress code.

After the ISU 100 has received a match to its unique identifier, the ISUwill receive the upstream frequency band command with a valid PINaddress code that tells the ISU 100 which 6 MHz band to use for upstreamtransmission and the carrier or tone designations for the upstream IOCchannel to be used by the ISU 100. The CXSU controller 102 interpretsthe command and correctly activates the ISU modem 101 of the ISU 100 forthe correct upstream frequency band to respond in. Once the ISU modem101 has acquired the correct 6 MHz band, the CXSU controller 102 sends amessage command to the ISU modem 101 to enable upstream transmission.Distributed loops utilizing the carrier, amplitude, and timing recoveryblock 222 of the MCC modem upstream receiver architecture of the HDT 12is used to lock the various ISU parameters for upstream transmission,including amplitude, carrier frequency, symbol alignment, and pathdelay.

The HDT is then given information on the new ISU 2712 and providesdownstream commands 2714 for the various parameters to the subscriberISU unit. The ISU begins transmission in the upstream 2720 and the HDT12 locks to the upstream signal 2730. The HDT 12 derives an errorindicator 2732 with regard to the parameter being adjusted and commands2734 the subscriber ISU to adjust such parameter. The adjustment oferror is repeated 2736 in the process until the parameter for ISUtransmission is locked to the HDT 12.

More specifically, after the ISU 100 has acquired the 6 MHz band foroperation, the CXSU 102 sends a message command to the ISU modem 101 andthe ISU modem 101 transmits a synchronization pattern on asynchronization channel in the primary synchronization band of thespectral allocation as shown in FIGS. 13-18. The upstreamsynchronization channels which are offset from the payload data channelsas allocated in FIGS. 13-18 include both a primary and a redundantsynchronization channel such that upstream synchronization can still beaccomplished if one of the synchronization channels is corrupted. TheHDT monitors one channel for every ISU.

The MCC modem 82 detects a valid signal and performs an amplitude levelmeasurement on a received signal from the ISU. The synchronizationpattern indicates to the CXMC 80 that the ISU 100 has received theactivation and initialization and frequency band commands and is readyto proceed with upstream synchronization. The amplitude level iscompared to a desired reference level. The CXMC 80 determines whether ornot the transmit level of the ISU 100 should be adjusted and the amountof such adjustment. If level adjustment is required, the CXMC 80transmits a message on the downstream IOC channel instructing the CXSU102 of the ISU 100 to adjust the power level of the transmitter of theISU modem 101. The CXMC 80 continues to check the receive power levelfrom the ISU 100 and provides adjustment commands to the ISU 100 untilthe level transmitted by the ISU 100 is acceptable. The amplitude isadjusted at the ISU as previously discussed. If amplitude equilibrium isnot reached within a certain number of iterations of amplitudeadjustment or if a signal presence is never detected utilizing theprimary synchronization channel then the same process is used on theredundant synchronization channel. If amplitude equilibrium is notreached within a certain number of iterations of amplitude adjustment orif a signal presence is never detected utilizing the primary orredundant synchronization channels then the ISU is reset.

Once transmission level adjustment of the ISU 100 is completed and hasbeen stabilized, the CXMC 80 and MCC modem 82 perform carrier frequencylocking. The MCC modem 82 detects the carrier frequency as transmittedby the ISU 100 and performs a correlation on the received transmissionfrom the ISU to calculate a carrier frequency error correction necessaryto orthogonally align the multicarriers of all the upstreamtransmissions from the ISUs. The MCC modem 82 returns a message to theCXMC 80 indicating the amount of carrier frequency error adjustmentrequired to perform frequency alignment for the ISU. The CXMC 80transmits a message on a downstream IOC channel via the MCC modem 82instructing the CXSU 102 to adjust the transmit frequency of the ISUmodem 101 and the process is repeated until the frequency has beenestablished to within a certain tolerance for the OFDM channel spacing.Such adjustment would be made via at least the synthesizer block 195(FIGS. 24 and 25). If frequency locking and adjustment is accomplishedon the ISU as previously described, then this frequency adjustmentmethod is not utilized.

To establish orthogonality, the CXMC 80 and MCC modem 82 perform symbolalignment. The MCC modem 82 detects the synchronization channelmodulated at a 8 kHz frame rate transmitted by the ISU modem 101 andperforms a hardware correlation on the receive signal to calculate thedelay correction necessary to symbol align the upstream ISU transmissionfrom all the distinct ISUs 100. The MCC modem 82 returns a message tothe CXMC 80 indicating the amount of delay adjustment required to symbolalign the ISU 100 such that all the symbols are received at the HDT 12simultaneously. The CXMC 80 transmits a message in a downstream IOCchannel by the MCC modem 82 instructing the CXSU controller 102 toadjust the delay of the ISU modem 101 transmission and the processrepeats until ISU symbol alignment is achieved. Such symbol alignmentwould be adjusted via at least the clock delay 196 (FIGS. 24 and 25).Numerous iterations may be necessary to reach symbol alignmentequilibrium and if it is not reached within a predetermined number ofiterations, then the ISU may again be reset.

Simultaneously with symbol alignment, the CXMC 80 transmits a message tothe MCC modem 82 to perform path delay adjustment. The CXMC 80 sends amessage on a downstream IOC channel via the MCC modem 82 instructing theCXSU controller 102 to enable the ISU modem 101 to transmit anothersignal on a synchronization channel which indicates the multiframe (2kHz) alignment of the ISU 100. The MCC modem 82 detects this multiframealignment pattern and performs a hardware correlation on the pattern.From this correlation, the modem 82 calculates the additional symbolperiods required to meet the round trip path delay of the communicationsystem. The MCC modem 82 then returns a message to the CXMC 80indicating the additional amount of delay which must be added to meetthe overall path delay requirements and the CXMC then transmits amessage on a downstream IOC channel via the MCC modem 82 instructing theCXSU controller 102 to relay a message to the ISU modem 101 containingthe path delay adjustment value. Numerous iterations may be necessary toreach path delay equilibrium and if it is not reached within apredetermined number of iterations, then the ISU may again be reset.Such adjustment is made in the ISU transmitter as can be seen in thedisplay delay buffer “n” samples 192 of the upstream transmitterarchitectures of FIGS. 24 and 25. Path delay and symbol alignment may beperformed at the same time, separately or together using the same ordifferent signals sent on the synchronization channel.

Until the ISU is initialized and activated, the ISU 100 has nocapability of transmitting telephony data information on any of the 480tones or carriers. After such initialization and activation has beencompleted, the ISUs are within tolerance required for transmissionwithin the OFDM waveform and the ISU is informed that transmission ispossible and upstream synchronization is complete.

After an ISU 100 is initialized and activated for the system, follow-upsynchronization or tracking may be performed periodically to keep theISUs calibrated within the required tolerance of the OFDM transportrequirements. The follow-up process is implemented to account for driftof component values with temperature. If an ISU 100 is inactive forextreme periods of time, the ISU can be tuned to the synchronizationchannels and requested to update upstream synchronization parameters inaccordance with the upstream synchronization process described above.Alternatively, if an ISU has been used recently, the follow-upsynchronization or tracking can proceed over an IOC channel. Under thisscenario, as generally shown in FIG. 28, the ISU 100 is requested toprovide a signal over an IOC channel by the HDT 12,2800. The HDT 12 thenacquires and verifies that the signal is within the tolerance requiredfor a channel within the OFDM waveform 2811. If not, then the ISU isrequested to adjust such errored parameters 2813. In addition, duringlong periods of use, ISUs can also be requested by the HDT 12 to send asignal on an IOC channel or a synchronization channel for the purpose ofupdating the upstream synchronization parameters.

In the downstream direction, the IOC channels transport controlinformation to the ISUs 100. The modulation format is preferablydifferentially encoded BPSK, although the differential aspect of thedownstream modulation is not required. In the upstream direction, theIOC channels transport control information to the HDT 12. The IOCchannels are differentially BPSK modulated to mitigate the transienttime associated with the equalizer when sending data in the upstreamdirection. Control data is slotted on a byte boundary (500 ps frame).Data from any ISU can be transmitted on an IOC channel asynchronously;therefore, there is the potential for collisions to occur. As there ispotential for collisions, detection of collisions on the upstream IOCchannels is accomplished at a data protocol level. The protocol forhandling such collisions may, for example, include exponential back-offby the ISUs. As such, when the HDT 12 detects an error in transmission,a retransmission command is broadcast to all the ISUs such that the ISUsretransmit the upstream signal on the IOC channel after waiting aparticular time; the wait time period being based on an exponentialfunction.

One skilled in the art will recognize that upstream synchronization canbe implemented allowing for multi-point to point transmission using onlythe symbol timing loop for adjustment of symbol timing by the ISUs ascommanded by the HDT. The frequency loop for upstream synchronizationcan be eliminated with use of high quality local free runningoscillators in the ISUs that are not locked to the HDT. In addition, thelocal oscillators at the ISUs could be locked to an outside reference.The amplitude loop is not essential to achieve symbol alignment at theHDT.

In the process described above with respect to initialization andactivation, including upstream synchronization, if for some reasoncommunication is lost between a large number of ISUs 100 and the HDT 12,after a period of time these ISUs 100 will require initialization andactivation once again. Such a case may arise when a fiber is cut andusers of multiple ISUs 100 are left without service. As initializationand activation is described above, only one ISU 100 would be initializedand activated at one time. The time frame for initialization andactivation of multiple ISUs 100 in this manner is shown in FIG. 19.

In FIG. 19, each ISU 100 is initialized, as previously described, byidentification of the ISU and acquisition by the ISU of the 6 MHz bandfor downstream transmission during a scanning period TSCAN, which is thetime period needed for the ISU to scan all of the downstream bandslistening for a matching identification message. In one embodiment,TSCAN is equal to 6.16 seconds. Of course this time period is going tobe dependant on the number of bands scanned, the time period necessaryfor synchronizing on the downstream synchronization channels, and thetime required to acquire an IOC channel within the band.

Further, as shown in FIG. 19, after each ISU has acquired a downstreamand upstream 6 MHz band, upstream synchronization is then performedduring a time period TEQUAL•TEQUAL may be defined as the period in whichan ISU should have received all messages from the CXMC 80 finishing theupstream synchronization process as described above, with a reasonableamount of iterations to accomplish such synchronization. At the veryleast, this time period is the time period necessary to accomplishsymbol timing such that symbols received from various ISUs 100 at theHDT 12 are orthogonal. This time period would be increased if amplitude,frequency and path delay synchronization is also performed as describedabove. Therefore, the time period necessary to serially initialize andactivate twelve ISUs, T_(SERIAL), as shown in FIG. 19 would be equal to12T_(SCAN)+12T_(EQUAL).

With a burst identification process as shall be described with respectto FIG. 20, the time period for initializing and activating twelve ISUs100 can be substantially reduced. This results in more ISUs 100 beingactivated more quickly and more users once again serviced in a shorterperiod of time. In the process of burst identification as shown by thetiming diagram of FIG. 20, the identification and acquisition ofmultiple ISUs 100 is performed in parallel instead of being performedserially as described above.

The periodicity of the identification messages sent by the CXMC 80during initialization and acquisition, when performed during normaloperating conditions when ISUs 100 are serially initialized, is designedto present a light load of traffic on the IOC channel but yet to allow areasonable identification duration. This periodicity duration is, forexample, 50 milliseconds. For the system to be able to handle bothsituations, serial identification and burst identification, thisperiodicity is kept the same. However, in burst identification, the IOCchannel traffic load is not important because the service of all theISUs 100 receiving communication via one of the CXMCs 80 utilizing theIOC channels has been terminated such as by the cut fiber. Therefore,during burst identification the IOC channels can be loaded more heavilyand identification messages for multiple ISUs 100 utilizing such IOCchannels are transmitted on the IOC channels at the same periodicity asduring serial identification, but the phase for the identificationmessages is different for each ISU.

Due to the period, and utilization of the IOC channels for theidentification messages during burst identification, there is a limit onthe number of identification messages which may be transmitted duringone Tsc,6,N period. If the periodicity is 50 milliseconds and the use ofthe IOC channel for a single identification message is 4 milliseconds,only twelve ISUs 100 may be identified during one TSCAN period duringburst identification. As described below with further reference to FIG.20, if the number of ISUs 100 desired to be burst identified is greaterthan twelve, then multiple groups of burst identifications are seriallyperformed.

One skilled in the art will recognize that the specified numbers fortime periods are for illustration only and that the present invention isnot limited to such specified time periods. For example, the periodicitymay be 100 ms and the number of ISUs identified during burstidentification may be 24. Many different time periods may be specifiedwith corresponding changes made to the other times periods specified.Further, burst identification may be accomplished having a periodicitydifferent than that for serial identification.

As shown by the timing diagram of FIG. 19, a single burst initializationand activation of twelve inactive ISUs 100 can be accomplished in thetime period TBURST which is equal to _(TSCAN)+12TEQUAL. This is an11TscAN difference from the process carried out serially. During theTSCAN period, identification messages for all twelve ISUs 100 beinginitialized are transmitted on the IOC channels for a CXMC 80. Thetwelve identification messages are each sent once during each 50millisecond period. The phase of each message is however different. Forexample, identification message for ISUO may be sent at time 0 and thenagain at 50 ms, whereas the identification message for ISU1 may be sentat time 4 milliseconds and then again at 54 milliseconds and so forth.

After the ISUs 100 being initialized have been identified andacquisition of the downstream 6 MHz band has occurred during T_(SCAN),then upstream synchronization is performed in a serial manner withrespect to each ISU identified during TSCAN• The upstreamsynchronization for the ISUs is performed during the time period equalto 12T_(EQUAL). The CXMC 80 would start the upstream synchronizationprocess in the same manner as described above for each ISU identified ina serial manner. The CXMC 80 sends to the ISU the upstream transmissionband in which the ISU being synchronized is to transmit within andenables the upstream synchronization process to begin. The upstreamsynchronization process for an ISU has been described in detail above.If an upstream transmission band is not received and upstreamsynchronization is not enabled for an ISU during the 12T_(EQUAL) timeperiod, then the ISU is reset at the end of the 12 T_(EQUA)L period by aperiod of time equal to T_(SCAN)+12 T_(EQUAL) to possibly performupstream synchronization in the next 12 TEQUAL period. Once a burstidentification period, TBURST, is completed, the process may be startedover again in a second TBURST period as shown in FIG. 20 if additionalISUs 100 are to be initialized and activated.

FIGS. 47 and 48 describe a control loop distributed through the systemfor acquiring and tracking an ISU, according to another embodiment ofthe invention. Loop 3900 shows steps executed by the ISU 66 or 68 at theleft, and those executed by HDT 12 at the right. Messages between thesetwo units are shown as horizontal dashed lines; the IOC channels carrythese messages. Steps 3910 prepare the ISU to communicate with the HDT.Step 3911 reads a pre-stored internal table 3912 indicating frequenciesof the valid RF downstream bands, along with other information. Next,step 3913 tunes the ISUs' narrow-band receive modem to the center of thefirst 6 MHz band in table 3912. Step 3913 a then fine-tunes it to one ofthe two sync channels in that band; assume that this “primary” channelis the one at the lower end of the downstream part of the band shown inFIG. 13. Step 3914 acquires the amplitude and frequency of this synctone. Briefly, equalizer 172, FIG. 22 or 23, is adjusted to bring theout put of the FFT to about 12 dB below its upper limit. Recovery block166 measures the time for ten frames of the sync tone, and compares itto the 1.25 ms correct time interval; the frequency of synthesizer 157is then adjusted accordingly. Rough timing is satisfactory at thispoint, because the control messages used below are simple low-frequencybinary-keyed signals. If sync cannot be acquired, exit 3915 causes block3913 a to retune to a secondary sync tone of the same 6 MHz band, theupper downstream tone of FIG. 13. If the ISU also fails to synchronizeto the secondary tone, exit 3916 causes block 3913 to tune to the centerof the next 6 MHz band in table 3912 and attempt synchronization again.If all bands have been tried, block 3913 continues to cycle through thebands again. When step 3914 has acquired a lock, steps 3920 listen formessages from the HDT. Step 3921 reads an internal predeterminedmanufacturer's serial number which uniquely identifies that ISU, andcompresses it into a shorter, more convenient format. Step 3922fine-tunes the ISU modem to a designated primary subband, such assubband 0 in FIG. 16.

Concurrently, steps 3930 begin a search for the new ISU. Step 3931receives an operator “ranging” command to connect an ISU having aparticular serial number. Step 3932 then continuously broadcasts a “PINmessage” 3933 on all IOC channels; this message contains the compressedform of the ISU serial number and a shorter personal identificationnumber (PIN) by which that ISU will be known. Step 3923 in the ISUcontinuously receives all PIN messages, and attempts to match thetransmitted compressed serial number with the number from table 3922 a.If it fails to do so after a period of time, exit 3924 caused step 3922to retune to another designated subband, such as subband 23 in FIG. 13,and try again. If no appropriate PIN message is received on thesecondary subband, exit 3925 returns to step 3913 to switch to another 6MHz band. When step 3923 receives the correct PIN message, step 3926latches the PIN into the ISU to serve henceforth as its address withinthe system. In some implementations, the full serial number or otherunique identifier of the physical ISU could serve directly as anaddress. However, this number occupies many bytes; it would be wastefulto transmit it every time a message is addressed to the ISU, or even touse it for ranging. Its compressed form, two bytes long, serves as ahash code which is practical to transmit in the continuous messages3933. The PIN is only one byte long, since addresses need be unique onlywithin each 6 MHz band, and are practical to use for identifying the ISUwhenever the HDT needs to communicate with it. Message 3927 informs theHDT that the ISU will respond to its PIN.

Steps 3940 set up the upstream communications from the ISU to the HDT.After 3941 receives PIN confirmation 3927, step 3942 sends a designationof the upstream frequency band to the ISU as an IOC message 3943. Thisfrequency may have been specified by the operator in step 3931, or mayhave been generated by the HDT itself. Step 3944 tunes the ISU modem tothis 6 Mhz band, and returns a confirmation 3945. Step 3946 thenfine-tunes to the primary upstream sync channel of that band, such asthe lower one in FIG. 13. Step 3947 enables an HDT receiver on thedesignated band. Steps 3950 adjust the transmitted power of the ISU inthe upstream direction; in a multipoint-to-point-system, the powerlevels of all ISUs must track each other in order to ensureorthogonality of the entire signal received by the modem of FIG. 26.Step 3951 transmits a ranging tone at an initial power level from theISU on this sync channel, which is sometimes called a ranging channel.At the HDT, step 3952 measures the received power level, and block 3953sends an IOC message 3954, causing step 3955 at the ISU to adjust thepower of its transmitter 200, FIG. 24, if necessary. If this cannot bedone, message 3956 causes step 3946 to retune to a secondary rangingchannel, such as the higher upstream sync tone of FIG. 13, and causesthe step 3947 to enable the secondary channel at the HDT. If this loopalso fails, exit 3957 reports a hard failure to the system logic.

Blocks 3960 align the symbol or frame timing between the ISU and theHDT. Step 3961 measures the phase of the ISUs' ranging tone with respectto the sync tone that the HDT is sending to all ISUs at all times. FIG.11 shows this signal, labeled “8 kHz frame elk” in FIG. 24. Step 3962sends messages 3963 as necessary to cause the ISU modem to adjust thetiming of its ranging tone, in step 3964. When this has completed, block3965 detects whether or not groups of four frames are aligned correctlyas between HDT and ISU; this grouping delineates boundaries of IOCmessages, which are four frames long. The sync tones continually repeata differential BPSK pattern of 1010 0101 0101 0101 over a period of 16frames; that is, each bit occupies 125 microseconds, the duration of oneframe. Thus, the space between the fourth and fifth bit, and between thesixteenth and first, can mark the multiframe boundaries. If alignment isincorrect, step 3966 sends message 3967, causing the ISU to bump thephase of its “2 kHz superframe elk,” FIG. 24. If step 3961 or step 3965cannot reach the correct phase after a certain number of steps, failexits 3969 report a hard failure.

Steps 3980 complete the induction of the new ISU into the system. Step3981 turns off the ranging tone at the ISU, tells the HDT at 3982 thatit is locked in, and returns to the subband at which it was operating instep 3944. Step 3983 requests preliminary configuration or capabilitydata from the ISU, in message 3984, whereupon step 3985 reads aninternal table 3986 containing parameters indicating capabilities suchas whether the ISU can tune only odd or even channel numbers, and otherphysical limitations of that particular modem. When message 3987 hascommunicated these parameters, step 3988 selects a particular subband of10 or 130 payload channels (for an HISU or an MISU respectively).Message 3989 causes step 3990 to tune the ISU to the proper subband.Meanwhile, the HDT is acquiring an IOC data-link (IDL) channel, asdescribed herein below, at step 3991. Step 3992 then sends message 3993to the ISU, which reads the rest of the modem configuration andspecifications from table 3986 at step 3994, and transmits them over theIDL channel at 3995. The HDT stores pertinent information concerningthat ISU for future reference. The purpose of sending ISU data to theHDT is to accommodate various ISU models having greatly differingcapabilities, and to allow continued use of legacy ISU equipment whenthe HDT has been upgraded to include additional capabilities.

During and after the process of FIGS. 47 and 48, the ISU receivingmodem, FIG. 22 or 23, must track the acquired frequency and symboltiming of the HDT transmitting modem of FIG. 21. The practicalities of amulticarrier (MC) system impose requirements which are not obvious fromexperience with TDMA (time division multiple-access) and otherconventional forms of bidirectional multipoint networks and systems, norin point-to-multipoint (“broadcast”) multicarrier networks. In TDMA andsimilar systems, slight errors in frequency and timing, and largererrors in amplitude, can be compensated by tracking the received signal.In a broadcast MC system, all carriers are synchronized at the head end,and can again be tracked at the receiver. In a bidirectionalmultipoint-to-point multicarrier system, however, the HDT receiver mustsee all channels as though they had been generated by a single source,because the HDT decodes all channels in an entire 6 MHz band as a singleentity, with a single FFT. Even slight errors among the individual ISUsin their 10-channel and 130-channel subbands causes severe distortionand inter-symbol interference when the HDT FFT inverts the channels backinto symbol strings for multiple DSO channels. The errors to becontrolled are frequency, symbol timing, and signal amplitude. Amplitude(power level) in particular has been found to be much more stringentthan in previous systems.

At all times after the ISU receiving modem FIG. 22 or 23 has acquiredthe signal of HDT transmitting modem FIG. 21, it must track gradualchanges in frequency, phase, and symbol timing caused by drift and otherchanges in the plant. FIG. 49 shows a method 4000 for tracking thesechanges. Steps 4010, executed by unit 153, FIG. 22 or 155, FIG. 23,track the downstream power level to compensate for slow gain changes inthe plant. Step 4011 measures an average signal level by known methodsfrom the output of FFT 170 or 180. If it is correct, exit 4012 takes noaction. If the error is wrong but within a predetermined threshold ofthe correct value, block 4013 adjusts a coefficient of equalizer 172 toreturn the signal level to its nominal value. If the signal levelchanges more than a certain amount, or if it changes rapidly enough, thesignal has probably been lost entirely. In that case, exit 4014 exitstracking mode; it may reenter the ranging procedure of FIGS. 47 and 48,or it may merely signal an error to the HDT.

Steps 4020 track carrier frequency in unit 166, FIG. 22 or 23. An HISUmodem of FIG. 23 receives a subband having ten payload channels and oneIOC at the center, as shown in FIG. 16. When this narrowband modem istuned to the subband, the sync tones are no longer within its frequencyrange. Therefore, step 4021 measures the phase of the IOC channel withinthe currently tuned subband, rather than the sync tone used in FIGS. 47and 48. Step 4022 smoothes any phase error between the received carrierand the locally generated signal from generator 168, FIG. 22, to preventjitter. The frame clocks of both the receiving modem FIG. 23 and theupstream transmitter FIG. 24 use this clock; that is, the clocks arelocked together within the HISU modem. Step 4023 updates the frequencyof a local oscillator in RF synthesizer 157. It should be noted herethat the location of the IOC in the middle of the subband eliminates anyoffset phase error which otherwise must be compensated for. Steps 4020may be the same for an MISU modem, FIG. 22; this modem has a 130-channelbandwidth, as shown in FIG. 16. The wider bandwidth of this modemincludes multiple IOC tones for tracking. The modem may use one of them,preferably near the middle, or pairs of tones offset from the center ofthe subband.

Step 4030 tracks symbol timing. Step 4031 samples the frequency errorbetween the received symbols and the local 8 kHz symbol sampling clock.If the sampling frequencies differ by more than about 2 ppm between theHDT and the ISU, the synthesized tones progressively depart from theirbins at the receiving FFT until the equalizer can no longer track them.Step 4032 receives the sign of the sampling error from step 4031, andapplies a small 0.5 ppm correction to the frame frequency. Process 4000takes places in real time, in parallel with other processes.

After the upstream transmitting modem 101 portion shown in FIG. 24 hastuned to a subband in FIGS. 47 and 48, it and the upstream receivingmodem 82 portion of FIG. 26 must continue to track in amplitude,frequency, and timing. The use of multicarrier (MC) technology imposessome requirements which are not obvious from experience with TDMA(time-division multiple access) or other bidirectional multipointtechnologies, nor from point-to multipoint (broadcast) MC networks andsystems. In TDMA and similar systems, slight errors in frequency andtiming, and larger errors in signal amplitude, can be compensated bytracking the signal at the receiver. In broadcast MC systems, allcarriers are inherently synchronized at the head end, and can be trackedadequately at each receiver separately. In a bidirectionalmultipoint-to-point multicarrier plant, however, the head-end receiversees all channels as though they had been generated by a single source,because the HDT decodes all channels in an entire 6 MHz band as a singleorthogonal waveform, with a single FFT converter. Even slight errorsamong the various ISUs in their 10-channel and 130-channel subbands cancause severe distortion and inter-symbol interference when the FFT inmodem 82 portion of FIG. 26 converts the channels back into bit stringsfor multiple DSO channels. The parameters to be controlled arefrequency, symbol timing, and signal amplitude or power level. Frequencyand timing can be tracked in a manner similar to steps 4020 and 4030 ofFIGS. 47 and 48. Amplitude, however, has been found to be more criticalthan in previous systems.

FIG. 50 depicts a method 4100 for tracking changes in the upstreamchannel signal amplitude. After FIGS. 47 and 48 have brought the ISUtransmitting modem of FIG. 24 on line, its amplitude must be balancedwith that of all other ISUs in the system. Again, if different upstreamsubbands were received by different hardware, or in a TDMA fashion,where amplitude tracking could be particularized in frequency and/ortime, a conventional AGC circuit could track amplitude variationsadequately. In the embodiment described, power variations greater thanabout 0.25 dB from one subband to another cause a significant amount ofdistortion and inter-symbol interference. In a physical system of, say,a 20 km radius, variations in the upstream signal level at the head endmay vary by 20 dB or more for different ISU locations, and mayadditionally vary significantly over time because of temperaturedifferences, channel loading, component aging, and many other factors.Conventional methods cannot achieve both the wide dynamic range and thehigh resolution required for an MC bidirectional multipoint-to pointsystem.

The steps in the left column of FIG. 50 are performed within each ISU;the HDT performs the steps on the right. Step 4110 selects a number ofpayload channels for monitoring from table 4111. The channels mustinclude one channel from each separate ISU, but need not include morethan one. An MISU thus needs time in only one of its 130 payloadchannels, a very low overhead. A 10-channel HISU subband, however, mayrequire time in more than one of its 10 payload channels, becausemultiple HISUs can share the same subband. Of course, a powered-downISU, or one having no upstream payload channels provisioned to it, neednot participate in blocks 3940 of FIG. 47, because it does not transmitupstream at all. (It is alternatively possible to employ IOC channelsinstead of payload channels for this purpose. Although requiring lessoverhead, such use is generally much more complex to implement.)

After ranging procedure 3900 has acquired a correct initial power level,step 4120 performs a scan every 30 msec. for all the selected payloadchannels, as indicated by arrow 4121. Each ISU responds at 4130 bysending a message 4131 on its selected upstream channel when the scanreaches that channel. In step 4140, the HDT measures the received powerlevel from each ISU separately. If the signal amplitude is within acertain range of its previous value, then steps 4150 compensate for thevariation at the HDT. Step 4151 smoothes the errors over several scans,to prevent sudden jumps from a single glitch. Step 4152 then adjusts thecoefficients of equalizer 214 in the upstream receiving portion, FIG.26, of modem 82, FIG. 3, to compensate for the variation. This sequencecompensates for small, slow variations at a high resolution; theequalizer steps are small and very linear.

Step 4140 may detect that the HDT equalizer is far from nominal, nearthe end of its range, say, 4 dB up or down from nominal, for anequalizer having a ±5 dB range of 0.25 dB steps. This condition mightoccur for a large number of accumulated small errors, or it could becaused by a sudden, major change in the system, such as a break in anoptical fiber followed by an automatic switch to another fiber having adifferent length. Steps 4160 compensate for this condition at thetransmitting (ISU) end, rather than at the receiving (HDT) end. The HDTthen stops the channel scan at step 4161, and step 4162 sends anamplitude-error message 4163 over an IOC channel, addressed to theoffending ISU. The message specifies the amount and direction of thecompensation to be applied. The ISU applies this correction at step 4170to vary the power output of its transmitter 200, FIG. 24. Conventionaltransmitter power controls, such as a PIN attenuator diode 201 in poweramplifier 200, are typically relatively coarse and nonlinear, but theydo possess a wide range of adjustment. DAC 203 receives IOC messages tocontrol attenuator 201. To allow the head-end equalizer to track thechanging power more easily, step 4162 applies the correction over a longperiod of time, for example, 4-5 sec/dB; but, if the monitored channelis the only channel connected to that ISU, the entire correction can bemade in a single large step. Instead of controlling adjustment speed atthe ISU, the head end may send individual timed IOC messages formultiple partial corrections; the downside is increased message trafficon the IOC channels. ISUs may come online and be powered down at oddtimes. To prevent a feckless attempt at correction when an ISU ispowered down, or its signal has been lost for some other reason, step4140 further detects a condition of substantially zero power receivedfrom the ISU. In that event, step 4180 sets the head-end equalizer toits default value and keeps it there.

Thus, power-leveling blocks 3950 take advantage of the characteristicsof the system to adjust both ends in a way which achieves both highresolution and wide dynamic range. The digital control available in thehead-end equalizer provides precision and linearity in tracking slowchanges, and the analog control at the ISU provides a wide range, andstill allows the head end to track out inaccuracies caused by its coarseand nonlinear nature.

Call processing in the communication system 10 entails the manner inwhich a subscriber is allocated channels of the system for telephonytransport from the HDT 12 to the ISUs 100. The present communicationsystem in accordance with the present invention is capable of supportingboth call processing techniques not involving concentration, forexample, TR-8 services, and those involving concentration, such asTR-303 services. Concentration occurs when there are more ISUterminations requiring service than there are channels to service suchISUs. For example, there may be 1,000 customer line terminations for thesystem, with only 240 payload channels which can be allocated to provideservice to such customers. Where no concentration is required, such asfor TR-8 operation, channels within the 6 MHz spectrum are staticallyallocated. Therefore, only reallocation of channels shall be discussedfurther below with regard to channel monitoring.

On the other hand, for dynamically allocated channels to provideconcentration, such as for providing TR-303 services, the HDT 12supports on-demand allocation of channels for transport of telephonydata over the HFC distribution network 11. Such dynamic allocation ofchannels is accomplished utilizing the IOC channels for communicationbetween the HDT 12 and the ISUs 100. Channels are dynamically allocatedfor calls being received by a customer at an ISU 100, or for callsoriginated by a customer at an ISU 100. The CXMU 56 of HDT 12, aspreviously discussed, implements IOC channels which carry the callprocessing information between the HDT 12 and the ISUs 100. Inparticular, the following call processing messages exist on the IOCchannels. They include at least a line seizure or off-hook message fromthe ISU to the HDT; line idle or on-hook message from the ISU to theHDT; enable and disable line idle detection messages between the HDT andthe ISUs.

Call processing in the communication system 10 entails the manner inwhich a subscriber is allocated channels of the system for telephonytransport from the HDT 12 to the ISUs 100. The present communicationsystem in accordance with the present invention is capable of supportingboth call processing techniques not involving concentration, forexample, TR-8 services, and those involving concentration, such asTR-303 services. Concentration occurs when there are more ISUterminations requiring service than there are channels to service suchISUs. For example, there may be 1,000 customer line terminations for thesystem, with only 240 payload channels which can be allocated to provideservice to such customers. Where no concentration is required, such asfor TR-8 operation, channels within the 6 MHz spectrum are staticallyallocated. Therefore, only reallocation of channels shall be discussedfurther below with regard to channel monitoring.

On the other hand, for dynamically allocated channels to provideconcentration, such as for providing TR-303 services, the HDT 12supports on-demand allocation of channels for transport of telephonydata over the HFC distribution network 11. Such dynamic allocation ofchannels is accomplished utilizing the IOC channels for communicationbetween the HDT 12 and the ISUs 100. Channels are dynamically allocatedfor calls being received by a customer at an ISU 100, or for callsoriginated by a customer at an ISU 100. The CXMU 56 of HDT 12, aspreviously discussed, implements IOC channels which carry the callprocessing information between the HDT 12 and the ISUs 100. Inparticular, the following call processing messages exist on the IOCchannels. They include at least a line seizure or off-hook message fromthe ISU to the HDT; line idle or on-hook message from the ISU to theHDT; enable and disable line idle detection messages between the HDT andthe ISUs.

For a call to a subscriber on the HFC distribution network 11, the CTSU54 sends a message to the CXMU 56 associated with the subscriber linetermination and instructs the CXMU 56 to allocate a channel fortransport of the call over the HFC distribution network 11. The CXMU 56then inserts a command on the IOC channel to be received by the ISU 100to which the call is intended; the command providing the properinformation to the CXSU 102 to alert the ISU 100 as to the allocatedchannel. When a call is originated by a subscriber at the ISU side, eachISU 100 is responsible for monitoring the channel units for lineseizure. When line seizure is detected, the ISU 100 must communicatethis change along with the PIN address code for the originating line tothe CXMU 56 of the HDT 12 using the upstream IOC operation channel. Oncethe CXMU 56 correctly receives the line seizure message, the CXMU 56forwards this indication to the CTSU 54 which, in turn, provides thenecessary information to the switching network to set up the call. TheCTSU 54 checks the availability of channels and allocates a channel forthe call originated at the ISU 100. Once a channel is identified forcompleting the call from the ISU 100, the CXMU 56 allocates the channelover the downstream IOC channel to the ISU 100 requesting line seizure.When a subscriber returns on hook, an appropriate line idle message issent upstream to the HDT 12 which provides such information to the CTSU54 such that the channel can then be allocated again to support TR-303services.

Idle channel detection can further be accomplished in the modemutilizing another technique. After a subscriber at the ISU 100 hasterminated use of a data payload channel, the MCC modem 82 can make adetermination that the previously allocated channel is idle. Idledetection may be performed by utilizing the equalization process byequalizer 214 (FIG. 26) which examines the results of the FFT whichoutputs a complex (I and Q component) symbol value. An error iscalculated, as previously described herein with respect to equalization,which is used to update the equalizer coefficients. Typically, when theequalizer has acquired the signal and valid data is being detected, theerror will be small. In the event that the signal is terminated, theerror signal will increase, and this can be monitored by signal to noisemonitor 305 to determine the termination of the payload data channelused or channel idle status. This information can then be utilized forallocating idle channels when such operation of the system supportsconcentration. The equalization process can also be utilized todetermine whether an unallocated or allocated channel is being corruptedby ingress as shall be explained in further detail below with respect tochannel monitoring.

The telephony transport system may provide for channel protection fromingress in several manners. Narrowband ingress is a narrowband signalthat is coupled into the transmission from an external source. Theingress signal which is located within the OFDM waveform can potentiallytake the entire band offline. An ingress signal is (most likely) notorthogonal to the OFDM carriers, and under worst case conditions caninject interference into every OFDM carrier signal at a sufficient levelto corrupt almost every DSO+ to an extent that performance is degradedbelow a minimum bit error rate.

One method provides a digitally tunable notch filter which includes aninterference sensing algorithm for identifying the ingress location onthe frequency band. Once located, the filtering is updated to provide anarbitrary filter response to notch the ingress from the OFDM waveform.The filter would not be part of the basic modem operation but requiresthe identification of channels that are degraded in order to “tune” themout. The amount of channels lost as a result of the filtering would bedetermined in response to the bit error rate characteristics in afrequency region to determine how many channels the ingress actuallycorrupted.

Another approach as previously discussed with respect to the ingressfilter and FFT 112 of the MCC upstream receiver architecture of FIG. 26is the polyphase filter structure. The cost and power associated withthe filter are absorbed at the HDT 12, while supplying sufficientingress protection for the system. Thus, power consumption at the ISUs100 is not increased. The preferred filter structure involves twostaggered polyphase filters as previously discussed with respect toFIGS. 31 and 32 although use of one filter is clearly contemplated withloss of some channels. The filter/transform pair combines the filter anddemodulation process into a single step. Some of the features providedby polyphase filtering include the ability to protect the receive bandagainst narrowband ingress and allow for scalable bandwidth usage in theupstream transmission. With these approaches, if ingress renders somechannels unusable, the HDT 12 can command the ISUs to transmit upstreamon a different carrier frequency to avoid such ingress.

The above approaches for ingress protection, including at least the useof digital tunable notch filters and polyphase filters, are equallyapplicable to point to point systems utilizing multicarrier transport.For example, a single MISU transporting to a single HDT may use suchtechniques. In addition, uni-directional multi-point to point transportmay also utilize such techniques for ingress protection.

In addition, channel monitoring and allocation or reallocation basedthereon may also be used to avoid ingress. External variables canadversely affect the quality of a given channel. These variables arenumerous, and can range from electro-magnet interference to a physicalbreak in an optical fiber. A physical break in an optical fiber seversthe communication link and cannot be avoided by switching channels,however, a channel which is electrically interfered with can be avoideduntil the interference is gone. After the interference is gone thechannel could be used again.

Referring to FIG. 40, a channel monitoring method is used to detect andavoid use of corrupted channels. A channel monitor 296 is used toreceive events from board support software 298 and update a channelquality table 300 in a local database. The monitor 296 also sendsmessages to a fault isolator 302 and to channel allocator 304 forallocation or reallocation. The basic input to the channel monitor isparity errors which are available from hardware per the DSO+ channels;the DSO+ channels being 10-bit channels with one of the bits having aparity or data integrity bit inserted in the channel as previouslydiscussed. The parity error information on a particular channel is usedas raw data which is sampled and integrated over time to arrive at aquality status for that channel.

Parity errors are integrated using two time frames for each of thedifferent service types including POTS, ISDN, DDS, and DS1, to determinechannel status. The first integration routine is based on a shortintegration time of one second for all service types. The secondroutine, long integration, is service dependent, as bit error raterequirements for various services require differing integration timesand monitoring periods as seen in Table 4. These two methods aredescribed below.

Referring to FIGS. 41, 42, and 43, the basic short integration operationis described. When a parity error 5000 of a channel is detected by theCXMU 56, a parity interrupt is disabled by setting the interruptpriority level above that of the parity interrupt 5001 (FIG. 41). If amodem alarm is received which indicates a received signal failure,parity errors will be ignored until the failure condition ends 5002.Thus, some failure conditions will supersede parity error monitoring.Such alarm conditions may include loss of signal, modem failure, andloss of synchronization. If a modem alarm is not active 5004, a paritycount table is updated 5006 and an error timer event as shown in FIG. 42is enabled (5008).

When the error timer event is enabled 5100, the channel monitor 296enters a mode wherein parity error registers of the CXMU 56 are readevery 10 milliseconds and error counts are summarized after a one secondmonitoring period 5105. Generally, the error counts are used to updatethe channel quality database 5334 and determine which (if any) channelsrequire re-allocation. The channel quality table 300 of the databasecontains an ongoing record of each channel. The table organizes thehistory of the channels in categories such as: current ISU assigned tothe channel, start of monitoring, end of monitoring, total error, errorsin last day, in last week and in last 30 days, number of seconds sincelast error, severe errors in last day, in last week and in last 30 days,and current service type, such as ISDN, assigned to the channel.

As indicated in FIG. 41, after the parity interrupt is disabled and noactive alarm exists, the parity counts are updated 5006 and the timerevent is enabled 5008. The timer event (FIG. 42), as indicated above,includes a one second loop where the errors are monitored. As shown inFIG. 42, if the one second loop has not elapsed 5110, the error countsare continued to be updated 5104. When the second has elapsed 5106, theerrors are summarized 5120. If the summarized errors over the one secondperiod exceed an allowed amount indicating that an allocated channel iscorrupted or bad 5121, as described below, channel allocator 304 isnotified 5123 and ISU transmission is reallocated to a differentchannel. As shown in FIG. 43, when the reallocation has been completed5200, the interrupt priority is lowered below parity 5210 so thatchannel monitoring continues and the channel quality database is updated5215 concerning the actions taken. The reallocation task may beaccomplished as a separate task from the error timer task or performedin conjunction with that task. For example, the reallocator 304 may bepart of channel monitor 296.

As shown in FIG. 44 in an alternate embodiment of the error timer task5110-2 of FIG. 42, channels can be determined to be bad 5304 before theone second has elapsed. This allows the channels that are determined tobe corrupted during the initial portion of a one second interval to bequickly identified and reallocated 5308 without waiting for the entireone second to elapse.

Instead of reallocation, the power level for transmission by the ISU maybe increased to overcome the ingress on the channel as indicated in FIG.124. However, if the power level on one channel is increased, the powerlevel of at least one other channel must be decreased as the overallpower level must be kept substantially constant. If all channels aredetermined bad 5306, the fault isolator 302 is notified 5320 indicatingthe probability that a critical failure is present, such as a fiberbreak. If the summarized errors over the one second period do not exceedan allowed amount indicating that the allocated channel is notcorrupted, the interrupt priority is lowered below parity 5210 and theerror timer event is disabled 5332. Such event is then ended 5350 andthe channels once again are monitored for parity errors per FIG. 41.

Two issues presented by periodic parity monitoring as described abovemust be addressed in order to estimate the bit error rate correspondingto the observed count of parity errors 1240 in a monitoring period ofone second to determine if a channel is corrupted. The first is thenature of parity itself. Accepted practice for data formats using blockerror detection assumes that a block error represents one bit of error,even though the error actually represents a large number of data bits.Due to the nature of the data transport system, errors injected intomodulated data are expected to randomize the data. This means that theaverage block error frame will consist of four (4) block error data bits(excluding the ninth bit). Since parity detects only odd bit errors,half of all block error frames are not detected by parity. Therefore,each parity (frame) error induced by transport interference representsan average of 8 (data) bits of error. Second, each monitoring parityerror represents 80 frames of data (10 ms/125 us). Since the parityerror is latched, all errors will be detected, but multiple errors willbe detected as one error.

The bit error rate (BER) used as a basis for determining when toreallocate a channel has been chosen as 10-3. Therefore, the acceptablenumber of parity errors in a one second interval that do not exceed 10-3must be determined. To establish the acceptable parity errors, theprobable number of frame errors represented by each observed (monitored)parity error must be predicted. Given the number of monitored parityerrors, the probable number of frame errors per monitored parity error,and the number of bit errors represented by a frame (parity) error, aprobable bit error rate can be derived 1242.

A statistical technique is used and the following assumptions are made:

-   -   1. Errors have a Poisson distribution, and    -   2. If the number of monitored parity errors is small (<10) with        respect to the total number of “samples” (100), the monitored        parity error rate (PER) reflects the mean frame error rate        (FER).

Since a monitored parity error (MPE) represents 80 frames, assumption 2implies that the number of frame errors (FEs) “behind” each parity erroris equal to 80 PER. That is, for 100 parity samples at 10 ms per sample,the mean number of frame errors per parity error is equal to 0.8 timesthe count of MPEs in one second. For example, if 3 MPEs are observed ina one second period, the mean number of FEs for each MPE is 2.4.Multiplying the desired bit error rate times the sample size anddividing by the bit errors per frame error yields the equivalent numberof frame errors in the sample. The number of FEs is also equal to theproduct of the number of MPEs and the number of FEs per MPE. Given thedesired BER, a solution set is determined as illustrated below inEquation 1.

$\begin{matrix}{\left( {{MPE}\;\frac{FE}{MPE}} \right) = {0.8\mspace{14mu}{MPE}}} & \left( {{Equation}\mspace{14mu} 1} \right)\end{matrix}$

The Poisson distribution, as illustrated below in Equation 2, is used tocompute the probability of a given number of FEs represented by a MPE(X), and assumption 2, above, is used to arrive at the mean number ofFEs per MPE (p).

$\begin{matrix}{{P(X)} = \frac{e - {\mu\;\mu\; x}}{x!}} & \left( {{Equation}\mspace{14mu} 2} \right)\end{matrix}$

Since the desired bit error rate is a maximum, the Poisson equation isapplied successively with values for x of 0 up to the maximum number.The sum of these probabilities is the probability that no more than xframe errors occurred for each monitored parity error.

The results for a bit error rate of 10⁻³ and bit errors per frame errorof 1 and 8 are shown below in Table 3.

TABLE 3 Bit Error Rate Probability Maximum Average Bit Errors MonitoredFrame Errors/ Frame Errors/ Probability pre Frame Parity MonitoredMonitored of Error Errors Parity Error (X) Parity Error (μ) BER <−10⁻³ 82 4 1.2 98% 3 3 2.4 78% 4 2 3.2 38% 1 8 8 6.4 80% 9 7 7.2 56% 10 7 8.045%

Using this technique, a value of 4 monitored parity errors detectedduring a one second integration was determined as the threshold toreallocate service of an ISU to a new channel. This result is arrived atby assuming a worst case of 8 bit errors per frame error, but aprobability of only 38% that the bit error rate is better than 10⁻³. Theproduct of the bit errors per frame, monitored parity errors and maximumframe errors per monitored parity error must be 64, for a bit error rateof 10⁻³ (64 errors in 64 k bits). Therefore, when the sampling of theparity errors in the error timer event is four or greater, the channelallocator is notified of a corrupted channel 1244. If the sampledmonitored parity errors is less than 4, the interrupt priority islowered below parity and the error timer event is disabled, ending thetimer error event and the channels are then monitored as shown in FIG.41.

The following is a description of the long integration operationperformed by the background monitor routine (FIG. 45) of the channelmonitor 296. The background monitor routine is used to ensure qualityintegrity for channels requiring greater quality than the shortintegration 10⁻³ bit error rate. As the flow diagram shows in FIG. 45,the background monitor routine operates over a specified time for eachservice type, updates the channel quality database 6006 table 300,clears the background count 6008, determines if the integrated errorsexceed the allowable limits determined for each service type 6010, andnotifies the channel allocator 304 of bad channels as needed 6012.

In operation, on one second intervals, the background monitor updatesthe channel quality database 6006 table. Updating the channel qualitydata table has two purposes. The first purpose is to adjust the biterror rate and number of errored seconds data of error free channels toreflect their increasing quality 1246. The second purpose is tointegrate intermittent errors on monitored channels which areexperiencing error levels too low to result in short integration timereallocation (less than 4 parity errors/second). Channels in thiscategory have their BER and numbers of errored seconds data adjusted,and based on the data, may be reallocated. This is known as longintegration time reallocation, and the default criteria for longintegration time re-allocation for each service type are shown asfollows in Table 4:

TABLE 4 Maximum Integration Monitory Service type: BER: Time: Erroredseconds Period: POTS 10⁻³ 1 second ISDN 10⁻⁶ 157 seconds   8% 1 hour DDS10⁻⁷ 157 seconds  0.5% 1 hour DS1 10⁻⁹ 15,625 0.04%  7 hours

Because POTS service does not require higher quality than 10⁻³,corrupted channels are sufficiently eliminated using the shortintegration technique and long integration is not required. As oneexample of long integration for a service type, the background monitorshall be described with reference to a channel used for ISDN transport.Maximum bit error rate for the channel may be 10⁻⁶, the number ofseconds utilized for integration time is 157, the maximum number oferrored seconds allowable is 8% of the 157 seconds, and the monitoringperiod is one hour. Therefore, if the summation of errored seconds isgreater than 8% over the 157 seconds in anyone hour monitoring period,the channel allocator 304 is notified of a bad channel for ISDNtransport.

Unallocated or unused channels, but initialized and activated, whetherused for reallocation for non-concentration services such as TR-8 orused for allocation or reallocation for concentration services such asTR-303, must also be monitored to insure that they are not bad, therebyreducing the chance that a bad channel will be allocated or reallocatedto an ISU 100. To monitor unallocated channels, channel monitor 296 usesa backup manager routine (FIG. 46) to set up unallocated channels in aloop in order to accumulate error data used to make allocation orre-allocation decisions. When an unallocated channel experiences errors6110, it will not be allocated to an ISU 100 for one hour 6118. Afterthe channel has remained idle (unallocated) for one hour, the channelmonitor places the channel in a loop back mode 6120 to see if thechannel has improved. In loop back mode, the CXMU 56 commands aninitialized and activated ISU 100 to transmit a message on the channellong enough to perform short or long integration on the parity errors asappropriate. In the loop back mode, it can be determined whether thepreviously corrupted channel has improved over time and the channelquality database is updated accordingly. When not in the loop back mode,such channels can be powered down. As described above, the channelquality database includes information to allow a reallocation orallocation to be made in such a manner that the channel used forallocation or reallocation is not corrupted. In addition, theinformation of the channel quality database can be utilized to rank theunallocated channels as for quality such that they can be allocatedeffectively. For example, a channel may be good enough for POTS and notgood enough for ISDN. Another additional channel may be good enough forboth. The additional channel may be held for ISDN transmission and notused for POTS. In addition, a particular standby channel of very goodquality may be set aside such that when ingress is considerably high,one channel is always available to be switched to.

In addition, an estimate of signal to noise ratio can also be determinedfor both unallocated and allocated channels utilizing the equalizer 214of the MCC modem 82 upstream receiver architecture as shown in FIG. 26and FIG. 125. As described earlier, the equalizer was previouslyutilized to determine whether a channel was idle such that it could beallocated. During operation of the equalizer, as previously described,an error is generated to update the equalizer coefficients. Themagnitude of the error can be mapped into an estimate of signal to noiseratio (SNR) by signal to noise monitor 305 (FIG. 26). Likewise, anunused channel should have no signal in the band. Therefore, by lookingat the variance of the detected signal within the unused FFT bin, anestimate of signal to noise ratio can be determined. As the signal tonoise ratio estimate is directly related to a probable bit error rate,such probable bit error rate can be utilized for channel monitoring inorder to determine whether a bad or good channel exists.

Therefore, for reallocation for non-concentration services such as TR-8services, reallocation can be performed to unallocated channels withsuch unallocated channels monitored through the loopback mode or by SNRestimation by utilization of the equalizer. Likewise, allocation orreallocation for concentration services such as TR303 services can bemade to unallocated channels based upon the quality of such unallocatedchannels as determined by the SNR estimation by use of the equalizer.

With respect to channel allocation, a channel allocator routine forchannel allocator 304 examines the channel quality database table todetermine which DSO+ channels to allocate to an ISU 100 for a requestedservice. The channel allocator also checks the status of the ISU andchannel units to verify in-service status and proper type for therequested service. The channel allocator attempts to maintain an optimaldistribution of the bandwidth at the ISUs to permit flexibility forchannel reallocation. Since it is preferred that ISUs 100, at leastHISUs, be able to access only a portion of the RF band at any giventime, the channel allocator must distribute channel usage over the ISUsso as to not overload anyone section of bandwidth and avoid reallocatingin-service channels to make room for additional channels.

The process used by the channel allocator 304 is to allocate equalnumbers of each ISU type to each band of channels of the 6 MHz spectrum.If necessary, in use channels on an ISU can be moved to a new band, ifthe current ISU band is full and a new service is assigned to the ISU.Likewise, if a channel used by an ISU in one band gets corrupted, theISU can be reallocated to a channel in another subband or band ofchannels. Remember that the distributed IOC channels continue to allowcommunication between the HDT 12 and the HISUs as an HISU always seesone of the IOC channels distributed throughout the spectrum. Generally,channels with the longest low-error rate history will be used first. Inthis way, channels which have been marked bad and subsequentlyreallocated for monitoring purposes will be used last, since theirhistories will be shorter than channels which have been operating in alow error condition for a longer period.

Second Embodiment of Telephony Transport System

A second embodiment of an OFDM telephony transport system, referring toFIGS. 36-39 shall be described. The 6 MHz spectrum allocation is shownin FIG. 36. The 6 MHz bandwidth is divided into nine channel bandscorresponding to the nine individual modems 226 (FIG. 37). It will berecognized by one skilled in the art that less modems could be used bycombining identical operations. Each of the channel bands includes 32channels modulated with a quadrature 32-ary format (32-QAM) having fivebits per symbol. A single channel is allocated to support transfer ofoperations and control data (IOC control data) for communication betweenan HDT 12 and ISUs 100. This channel uses BPSK modulation.

The transport architecture shall first be described with respect todownstream transmission and then with respect to upstream transmission.Referring to FIG. 37, the MCC modem 82 architecture of the HDT 12 willbe described. In the downstream direction, serial telephony informationand control data is applied from the CXMC 80 through the serialinterface 236. The serial data is de-multiplexed by demultiplexer 238into parallel data streams. These data streams are submitted to a bankof 32 channel modems 226 which perform symbol mapping and FFT functions.The 32 channel modems output time domain samples which pass through aset of mixers 240 that are driven by the synthesizer 230. The mixers 240create a set of frequency bands that are orthogonal, and each band isthen filtered through the filter/combiner 228. The aggregate output ofthe filter/combiner 228 is then up-converted by synthesizer 242 andmixer 241 to the final transmitter frequency. The signal is thenfiltered by filter 232, amplified by amplifier 234, and filtered againby filter 232 to take off any noise content. The signal is then coupledonto the HFC distribution network via telephony transmitter 14.

On the downstream end of the HFC distribution network 11, an ISU 100includes a subscriber modem 258 as shown in FIG. 38. The downstreamsignals are received from the ODN 18 through the coax leg 30, and arefiltered by filter 260 which provides selectivity for the entire 6 MHzband. Then the signal is split into two parts. The first part providescontrol data and timing information to synchronize clocks for thesystem. The second part provides the telephony data. With the controldata received separately from the telephony data, this is referred to asan out of band ISU (as previously described above). The out of bandcontrol channel which is BPSK modulated is split off and mixed tobaseband by mixer 262. The signal is then filtered by filter 263 andpassed through an automatic gain control stage 264 and a Costas loop 266where carrier phase is recovered. The signal that results is passed intoa timing loop 268 so timing can be recovered for the entire modem. TheIOC control data, which is a byproduct of the Costas loop, is passedinto the 32 channel OFDM modem 224 of the ISU 100. The second part ofthe downstream OFDM waveform is mixed to base band by mixer 270 andassociated synthesizer 272. The output of the mixer 270 is filtered byfilter 273 and goes through a gain control stage 274 to prepare it forreception. It then passes into the 32 channel OFDM modem 224.

Referring to FIG. 39, the IOC control data is hard limited throughfunction block 276 and provided to microprocessor 225. The OFDMtelephony data is passed through an analog to digital converter 278 andinput to a first-in first-out buffer 280 where it is stored. When asufficient amount of information is stored, it is accepted by themicroprocessor 225 where the remainder of the demodulation process,including application of an FFT, takes place. The microprocessor 225provides the received data to the rest of the system through the receivedata and receive data clock interface. The FFT engine 282 is implementedoff the microprocessor. However, one skilled in the art will recognizethat the FFT 282 could be done by the microprocessor 225.

In the upstream direction, data enters the 32 channel OFDM modem 224through the transmit data ports and is converted to symbols by themicroprocessor 225. These symbols pass through the FFT engine 282, andthe resulting time domain waveform, including guard samples, goesthrough a complex mixer 284. The complex mixer 284 mixes the waveform upin frequency and the signal is then passed through a random accessmemory digital to analog converter 286 (RAM DAC). The RAM DAC containssome RAM to store up samples before being applied to the analog portionof the ISU upstream transmitter (FIG. 38). Referring to FIG. 38, theOFDM output for upstream transport is filtered by filter 288. Thewaveform then passes through mixer 290 where it is mixed under controlof synthesizer 291 up to the transmit frequency. The signal is thenpassed through a processor gain control 292 so that amplitude levelingcan take place in the upstream path. The upstream signal is finallypassed through a 6 MHz filter 294 as a final selectivity before upstreamtransmission on the coaxial leg 30 to the ODN 18.

In the upstream direction at the HDT 12 side, a signal received on thecoax from the telephony receiver 16 is filtered by filter 244 andamplified by amplifier 246. The received signal, which is orthogonallyfrequency division multiplexed, is mixed to baseband by bank of mixers248 and associated synthesizer 250. Each output of the mixers 248 isthen filtered by baseband filter bank 252 and each output time domainwaveform is sent then to a demodulator of the 32 channel OFDM modems226. The signals pass through a FFT and the symbols are mapped back intobits. The bits are then multiplexed together by multiplexer 254 andapplied to CXMC 56 through the other serial interface 256.

As shown in this embodiment, the ISU is an out of band ISU asutilization of separate receivers for control data and telephony data isindicative thereof as previously discussed. In addition, the separationof the spectrum into channel bands is further shown. Various otherembodiments as contemplated by the accompanying claims of the transportsystem are possible by building on the embodiments described herein. Inone embodiment, an IOC control channel for at least synchronizationinformation transport, and the telephony service channels or paths areprovided into a single format. The IOC link between the HDT 12 and theISUs 100 may be implemented as four BPSK modulated carriers operating at16 kbps, yielding a data rate of 64 kbps in total. Each subscriber wouldimplement a simple separate transceiver, like in the second embodiment,which continuously monitors the service channel assigned to it on thedownstream link separately from the telephony channels. This transceiverwould require a tuned oscillator to tune to the service IOC channel.Likewise, an IOC channel could be provided for channel bands of the 6MHz bandwidth and the channel bands may include orthogonal carriers fortelephony data and an IOC channel that is received separately from thereception of the orthogonal carriers.

In another embodiment, instead of 4 BPSK channels, a single 64 kbps IOCchannel is provided. This single channel lies on the OFDM frequencystructure, although the symbol rate is not compatible with the telephonysymbol rate of OFDM framework. This single wide band signal requires awider band receiver at the ISU 100 such that the IOC link between theHDT 12 and ISUs is always possible. With single channel support it ispossible to use a fixed reference oscillator that does not have to tuneacross any part of the band in the subscriber units. However, unlike inthe first embodiment where the IOC channels are distributed across thespectrum allowing for narrow band receivers, the power requirements forthis embodiment would increase because of the use of the wide bandreceiver at the ISU 100.

In yet another embodiment, the IOC link may include two IOC channels ineach of 32 OFDM channel groups. This increases the number of OFDMcarriers to 34 from 32 in each group. Each channel group would consistof 34 OFDM channels and a channel band may contain 8 to 10 channelsgroups. This approach allows a narrow band receiver to be used to lockto the reference parameters provided by the HDT 12 to utilize an OFDMwaveform, but adds the complexity of also having to provide the controlor service information in the OFDM data path format. Because thesubscriber could tune to anyone of the channel groups, the informationthat is embedded in the extra carriers must also be tracked by thecentral office. Since the system needs to support a timing acquisitionrequirement, this embodiment may also require that a synchronizationsignal be place off the end of the OFDM waveform.

It is to be understood, however, that even though numerouscharacteristics of the present invention have been set forth in theforegoing description, together with details of the structure andfunction of the invention, the disclosure is illustrative and changes inmatters of order, shape, size, and arrangement of the parts, and variousproperties of the operation may be made within the principles of theinvention and to the full extent indicated by the broad general meaningof the terms in which the appended claims are expressed.

In the following detailed description of the preferred embodiments,reference is made to the accompanying drawings that form a part hereof,and in which are shown by way of illustration specific embodiments inwhich this invention may be practiced. It is understood that otherembodiments may be used and structural changes may be made withoutdeparting from the scope of the claimed invention.

FIG. 70 shows one embodiment of this invention having an apparatus,generally indicated as FFT system 2100, which performs both forward andinverse FFT functions. The input/output signals to FFT system 2100include some (or, in one preferred embodiment, all) of the following:real data-in 2111 (having an N-bit-wide data path; in one embodiment,this is 10 bits wide, and bi-directional so it can be both written toand read from), imaginary data-in 2112 (having an N-bit-wide data path;in one embodiment, this is 10 bits wide, and bi-directional so it can beboth written to and read from), address in 2113 (having enough bits tospecify addresses for each input value or input-value pair), control andclock lines 2114 that control writing (and reading) data from the inputside, test signals 2115, size-select bits 2116, bit-growth-select bitsfor each of P stages including bit-growth-1 bits 2117 throughbit-growth-P bits 2118, forward/inverse select signal 2119 whichspecifies whether to perform a forward or inverse transform, power-downcommand signal input 2109, real data-out 2121 (having an M-bit-wide datapath, in one embodiment, this is 10 bits), imaginary data-out 2122(having an M-bit-wide data path, in one embodiment, this is 10 bits),address out 2123 (having enough bits to specify addresses for eachoutput value or output value pair) (in one embodiment, these are drivenby an external device to select output values as they are needed),control and clock lines 2124 that control reading data from the outputside, overflow signal 2125 which indicates that one or more outputvalues has overflowed, and FFT complete signal 2126 which is activatedwhen a transform has been completed.

ASIC 2101 Fabrication

In one embodiment, the FFT system 2100 is fabricated on an applicationspecific integrated circuit (“ASIC”) 2101, a chip fabricated by LSILogic Inc. In this embodiment, full scan testing circuits are includedinto the ASIC 2101 for testability. In this embodiment, FFT system 2100is fabricated in LSI Logic LCB500K technology, which is a 0.5 micronrule, 3.3 Volt CMOS (complementary metal-oxide semiconductor) process.

Functional Description, Overview of FFT System 2100

In one embodiment, ASIC 2101 has four pins, size select 2116, to selectbetween the various transform sizes (that is, transforms having 2^(N)points; where 5N10, thus providing selectability for a 1024-pointtransform, a 512-point transform, a 256-point transform, a 128-pointtransform, a 64-point transform, or a 32-point transform). In oneapplication, transforms are completed in less than 125 microseconds. Seethe “Execution Time” section below for the minimum clock frequenciesnecessary to meet this requirement. In one embodiment, when performing a1024-point transform, a clock of at least approximately 32 MHz isrequired. In one embodiment, when performing a 512-point transform, aclock of at least approximately 16.5 MHz is required. In one embodiment,when performing a 32-point transform, a clock of at least approximately4 MHz is required.

FIG. 71 is a block diagram of modem 2400 according to the presentinvention which includes a FFT system 2100 configured to perform an IFFTin transmitter section 2401 (similar to the transmitter shown in FIG.21) and another FFT system 2100 configured to perform an FFT in receiversection 2402 (similar to the receiver shown in FIG. 26). In oneembodiment, ASIC 2101 has three logical banks of RAM which areconfigurable as shown in FIG. 72: an input RAM 2251 (containing aplurality of real input values 2241 and a plurality of imaginary inputvalues 2245), an output RAM 2253 (containing a plurality of real outputvalues 2243 and a plurality of imaginary output values 2247), and aconversion RAM 2252 (containing a plurality of real conversion values2242 and 2244 and a plurality of imaginary conversion values 2246 and2248). In one embodiment, input RAM 2251 has 1024 complex-valuepositions, each 20 bits wide (10 bits wide for each real input value2241 and 10 bits wide for each imaginary input value 2245), output RAM2253 has 1024 complex-value positions, each 20 bits wide (10 bits widefor each real output value 2243 and 10 bits for each imaginary outputvalue 2247) and conversion RAM 2252 has 1024 complex-value positions,each 30 bits wide (15 bits wide for each real conversion value 2242 and2244 and 15 bits wide for each imaginary conversion value 2246 and2248). (The 5 extra low-order real bits 2244 and 5 extra low-orderimaginary bits 2248 in each position of the conversion RAM 2252 help toavoid loss of precision during calculations.) The internal functions ofASIC 2101 have exclusive access to the conversion RAM 2252 and performthe FFT calculations out of this conversion RAM 2252. The input RAM 2251is accessible to the user's input device as an input to the FFT system2100 (to be written under external control with input data). The outputRAM 2253 is accessible to the user's output device as an output source(to be read under external control to obtain output data). In oneembodiment, the rising edge of an approximately 8-KHz frame clock 2108is used to start the computation of each transform (for example, eithera 1024-point FFT, or a 1024-point IFFT).

In one embodiment, the functions of all three banks of RAM are also“permuted” on the rising edge of this 8-KHz frame clock 2108, beforeeach FFT calculation starts. As used herein, this “permutation” changesthe function of each bank of RAM without actually moving data: input RAM2251 becomes conversion RAM 2252, conversion RAM 2252 becomes output RAM2253, and output RAM 2253 becomes input RAM 2251. Note that, in thisembodiment, no RAM data is moved when the functions are permuted. FIG.73 shows one embodiment of a physical implementation which provides thefunction of input RAM 2251, conversion RAM 2252, and output RAM 2253.Bank control block 2131 permutes the function of the physical RAM banks2151, 2152 and 2153 at the rising edge of frame clock 2139. One of thefunctions of bank control state machine 2131 is to control the routingof data through RAM input multiplexers (MUXs) 2171, 2172, and 2173 andthe routing of data through RAM output multiplexers (MUXs) 2181, 2182,and 2183. For example, at a first given state, bank control 2131controls input-select block 2132 to input data into physical RAM bank2151 (in one embodiment, physical RAM bank 2151 includes 1024 ten-bitreal values and 1024 ten-bit imaginary values). Thus, in the firststate, physical RAM bank 2151 acts as logical input RAM bank 2251. Onceall of the desired first set of input values (up to 1024 values or pairsof values) have been inputted, the frame clock 2139 is driven to changethe state of bank control 2131 (permuting the three RAM bank'srespective functions) to a second state, in which bank control 2131controls input-select block 2132 to input data into physical RAM bank2152 and bank control 2131 controls conversion-select block 2133 todirect computation accesses for data into physical RAM bank 2151. (Thefive low-order bits of values used for computation are always providedfrom physical RAM bank 2154.) Thus, in the second state, physical RAMbank 2152 acts as logical input RAM bank 2251 and physical RAM bank 2151acts as logical conversion RAM bank 2252.

Once all of the desired second set of input values (up to 1024 values orpairs of values) have been inputted, the frame clock 2139 is driven tochange the state of bank control 2131 (permuting the three RAM bank'srespective functions again) to a third state, in which bank control 2131controls input-select block 2132 to input data into physical RAM bank2153, bank control 2131 controls conversion-select block 2133 to directcomputation accesses for data into physical RAM bank 2152, and bankcontrol 2131 controls output-select block 2134 to direct output requestsfor accesses for data from physical RAM bank 2151. Thus, in the thirdstate, physical RAM bank 2153 acts as logical input RAM bank 2251,physical RAM bank 2152 acts as logical conversion RAM bank 2252, andphysical RAM bank 2151 acts as logical output RAM bank 2253. Once all ofthe desired third set of input values (up to 1024 values or pairs ofvalues) have been inputted, and the first set of converted output values(up to 1024 values or pairs of values) have been outputted, then theframe clock 2139 is driven to change the state of bank control 2131(permuting the three RAM bank's respective functions again) back to thefirst state, in which bank control 2131 controls input-select block 2132to input data into physical RAM bank 2151, bank control 2131 controlsconversion-select block 2133 to direct computation accesses for datainto physical RAM bank 2152, and bank control 2131 controlsoutput-select block 2134 to direct output requests for accesses for datafrom physical RAM bank 2153. Thus, in the first state, physical RAM bank2151 again acts as logical input RAM bank 2251, physical RAM bank 2153acts as logical conversion RAM bank 2252, and physical RAM bank 2152acts as logical output RAM bank 2253.

At the beginning of each transform (that is, each forward FFT or inverseFFT) process, the functions of the RAM bank memories (the mapping ofRAMs 2151, 2152, and 2153 to RAMs 2251, 2252, and 2253) are permuted.The conversion RAM 2252 becomes the output RAM 2253, the input RAM 2251becomes the conversion RAM 2252, and the output RAM 2253 becomes theinput RAM 2251. Each RAM (2251, 2252, and 2253) has its own independentcontrol and clock signals (2114, 2128, and 2124, respectively). Whilethe current transform is being calculated in conversion RAM 2252, theresults of the previously calculated FFT/IFFT may be read from theoutput RAM 2253 and the data for the next FFT calculation may beasynchronously and simultaneously written to the input RAM 2251. In oneembodiment, access to the input RAM 2251 and output RAM 2253 isrestricted for 3 clock cycles (one prior to and two after) relative tothe rising edge of the frame clock. This allows the RAM permutation toproceed safely without undesirable data loss. Forward Fast FourierTransform (FFT) and Inverse Fast Fourier Transform (IFFT).

In one embodiment, FFT system 2100 is implemented on a single integratedcircuit (K′) that performs both a forward FFT and an inverse FFT (IFFT,also called a reverse FFT). A pin, forward/inverse signal 2119, selectsbetween the two types of transform. The inverse FFT uses the identicalcalculation sequence as the forward transform, but the complex values(that is, real and imaginary) of the twiddle factors and butterflycoefficients are conjugated relative to these values used for the FFT. Aforward FFT is defined to convert a time-domain signal intofrequency-domain signals and, in one embodiment, is used in thetransmitter 2401 of modem 2400 (see FIG. 71). An inverse FFT is definedto convert frequency-domain signals to time-domain signals and is usedin the receiver 2402 of the modem 2400.

In one embodiment, the underlying structure of FFT system 2100 supportsfive radix-4 butterflies (that is, butterfly operations) usable toperform the 1024-point transforms (FFT and IFFT). The butterflies andstages are reduced for lower-order transforms. (In one embodiment, everyother twiddle factor of the 1024-point twiddle factor lookup table 2610is used when calculating the 512-point transforms by forcing to zero thelow-order address bit to the twiddle-factor lookup table 2610. Everyfourth twiddle factor of the 1024-point twiddle-factor lookup table 2610is used when calculating the 256-point transforms by forcing to zero thetwo low-order address bits to the twiddle-factor lookup table 2610.) Inone embodiment, all FFT transforms are calculated by using radix-4butterflies, except for the last stage of the 512-point, 128-point, and32-point transforms, which use the radix-4 structure to perform aradix-2 butterfly. (Obvious extensions are made if other transforms areused). Thus, the 1024 point transforms use five stages of radix-4butterflies, the 512-point transforms use four stages of radix-4butterflies followed by one stage of a radix-2 butterfly, etc.

Scaling of FFT Output

In one embodiment, scaling is controlled by ten external pins(bit-growth signals 2117 though 2118) on the ASIC 2101. Two pins areused for each of the five passes required for the 1024- and 512-pointtransform. The use of two pins at each stage allows a scaling factor(for example, a shift right of each intermediate result value) of from 0bits to 3 bits which exceeds the nominal bit growth of 1 bit observed inthe inventor's analysis. In one embodiment, the binary numberrepresented by each pair of pins indicates the number of binary placesthat the results of each calculation are to be shifted right (divided)by before they are placed back into conversion RAM 2252. Note that inone such embodiment scaling the result after each butterfly operationrequires at least some working registers to maintain bits greater thanthe MSB of the 15-bit values (each for real and imaginary) in theinternal conversion RAM 2252, and the scaling to be applied prior tostorage in the conversion RAM 2252.

Round-Off/Truncation, Saturation, and Scaling

In one embodiment, each input and output number is represented by 10bits (that is, 10 bits for each real portion, plus 10 bits for eachimaginary portion of a complex pair of numbers). Numbers are representedas two's-complement fractional arithmetic with the binary pointimmediately to the right of the sign bit. Fractional arithmetic helpsprevent multiplies from causing overflow. The result portion of adouble-precision multiply which is used is the upper fifteen bits.

Number growth is managed by the 10 scaling pins (bit growth signals2117-2118) mentioned previously. In addition, the ASIC 2101 usessaturation logic that prevents number roll over. That is to say, that ifthe result of an addition or subtraction exceeds the maximum value thatcan be represented in 15 bits, the result is replaced by a valuerepresenting the maximum possible integer. Likewise, if the result of anaddition or subtraction is less than the minimum value that can berepresented in 15 bits, the result is replaced by a value representingthe minimum possible integer.

In one embodiment, ASIC 2101 has the following registers for real andcomplex data: 10-bit input RAM 2251 and 10-bit output RAM 2253, and a 15bit conversion RAM 2252. The input values are presented as a two'scomplement fractional binary value (2.bbbbbbbbb). In the internalconversion RAM, the five extra bits are appended at the bottom (that is,low-order position) of the input word. The resulting format is anextended 2's-complement value in the internal conversion RAM 2252 (forexample, s.bbbbbbbbbxxxxx, where s is sign, b's are significant inputbits, and x's are extra bits to maintain precision). Note that in oneembodiment any extension bits in the working registers extend the signbit to achieve the desired 2's-complement results.

In one embodiment, at the output of each transform pass, the result isshifted down as specified by the signals provided into the scaling pins2117-2118 for that pass. The output RAM 2253 aligns precisely with theupper 10 bits of the conversion RAM 2252, in a manner similar to theinput RAM 2251 (s.bbbbbbbbbxxxxx), each output value (after the finalgrowth factor is applied prior to storage in the internal conversion RAM2252 is rounded to produce the output results (s.bbbbbbbbr).

Power Down

In one embodiment, the ASIC has a single pin, power-down signal 2109, tocontrol power down that gates off the internal clock in the ASIC, andplaces the input RAM 2251 and output RAM 2253 into a low power state. Inone embodiment, access attempts to those RAMS 2251 and 2253 while in thelow power state will be unsuccessful.

Execution Time

The maximum FFT execution frequency requires that a 1024-point transformbe completed in 125 microseconds based on a 32 MHz clock. Anapproximately 8-KHz frame clock 2108 is provided to the ASIC 2101 tosignify the beginning of a transform cycle. Prior to the rising edge ofthis clock, the previous FFT should be complete. Table 5 belowsummarizes the clock frequency to perform a 1024-point and a 512-pointFFT in 125 ms.

TABLE 5 FFT Clock required to perform a transform in 125 microsecondsClock cycles Clock Frequency for 125 μs Transform Size per conversionconversion rate (MHz) 1024 3845 30.8 512 1925 15.4 256 775 6.2 128 3903.1 64 150 1.2 32 80 0.64

FFT Functional Blocks

The ASIC is partitioned into the following major functional blocks.

-   -   1. RAM banks 2151, 2152, 2153    -   2. Sequencer 2640    -   3. Dual radix (2, 4) core 2600, including multipliers 2620        through 2627    -   4. Twiddle-factor lookup table 2610

The algorithm used is decimation in time (DIT) FFT. The algorithm usedis an in place algorithm which means that the results of each butterflyare put back into the same locations that the operands came from. Thealgorithm assumes digit reversed input order and normal output order.Since the three banks of RAMS are independent, the reordering of theinput data is done internally to the ASIC. This reordering istransparent to the user. Thus the user writes the data into the inputRAM bank in normal order and read the data from the output RAM bank innormal order. If the user requires some other ordering, this may beaccomplished by simply permuting the address line ordering. The order ofthe memory for all transform sizes are given in the section titled“Sequencer.”

Memory Banks

The ASIC 2101 includes three banks of RAM that are individuallyaddressable. The RAM banks 2251, 2253, and 2252 are used for input data,output data, and conversion data, respectively. The three banks are usedin order to obtain real-time execution of the FFT and to allowindependent clock rates for the input and output of data in order toaccommodate clocking needs of those devices connected to it. In oneembodiment, the input RAM bank 2251 and output RAM bank 2252 are eachorganized as 1024 twenty-bit words. The lower 10 bits of each 20-bitword are used for real data, and the upper 10 bits of each 20-bit wordare used for imaginary data. The conversion RAM bank 2252 is organizedas 1024 thirty-bit words. In one embodiment, the lower 15 bits of eachthirty-bit word are used for real data and the upper 15 bits forimaginary data. The connectivities of the three RAM banks of memory arecontrolled by a state machine which is advanced by the 8-KHz framesignal 2108. The state machine has three states which are defined inTable 6 as follows:

TABLE 6 Typical processing states of memory banks State BANK A BANK BBANK C 0 Input Convert Output 1 Convert Output Input 2 Output InputConvert

In one embodiment, the RAM banks do not have a reset. For input andoutput, they are accessed at a maximum of 10.24 MHz rate. In oneembodiment, a dead time of one clock cycle before the rising edge of the8-KHz clock 2108 and two clock cycles after is required to assure thesafety of the RAM bank switching. The ASIC 2101 clocks the data in onthe rising edge of the input clock and clocks data out on the risingedge of the output clock.

Actual operation of the conversion RAM 2252 is a bit more complicatedthan is implied above. Actually only the top 10 bits of the conversionRAM 2252 participates in the bank switching between RAMS 2151, 2152, and2153. The lower 5 bits 2154 are dedicated the conversion RAM 2252. Sinceall numbers are MSB aligned, no shifting needs to be done on input oroutput. However, the last stage of the FFT calculation rounds theresults to ten bits. This eliminates a bias that would result had theresults simply been truncated. The rounding is accomplished by adding aone to the eleventh bit position and then truncating this result.

Sequencer 264

The sequencer 2640 manages the processing of the FFT system 2100. Thesequencer 2640 controls the generation of addresses for the conversionRAM bank 2252 and the twiddle factor ROMs 2610 through addressgeneration blocks 2642 and 2641 respectively. Sequencer 2640 alsoconfigures the calculation commands for the radix butterfly calculator2630. In addition, sequencer 2640 monitors the calculations foroverflow. If at any time during the course of FFT calculation, anoverflow or underflow is detected, then a flag is set indicating thatthe results of the FFT are suspect. This overflow flag is passed alongwith the output data block when the RAM banks are switched. Thus theflag pin 2125 indicates that the output data block presently being readout of the output RAM bank 2253 may not be accurate. This flag 2125 mayaid the system designer in providing an indicator for AGC (automaticgain control).

Dual Radix (2, 4) Core 2600

The dual radix core 2600 is the arithmetic element of the ASIC 2101. Inone preferred embodiment, it includes eight 16-bit-by-16-bit multipliers(2620 through 2627) and thirty-two multiplexedadder-subtractor-accumulators 2633. In another embodiment, it includestwelve 16-bit-by-16-bit multipliers and eight multi-branchedadder/subtractor/accumulators.

FIG. 74 shows one embodiment of a dual radix core 2600. In theembodiment shown in FIG. 74, conversion RAM bank 2252 is shown as partof the dual radix core. In other embodiments, conversion RAM bank 2252is a separate functional unit, not considered part of the dual radixcore 2600. In FIG. 74, the data fetched from conversion RAM bank 2252 isfetched into holding latches 2612, with both the real (that is, X3R,X2R, X1R, and XOR) and imaginary parts (that is, X3I, X2I, XII, and X0I)of four points being fetched in parallel, substantially simultaneously.As shown in FIG. 74, the real and imaginary data fetched from conversionRAM bank 2252 is addressed by address generation circuit 2642 whichprovides four separate addresses to conversion RAM bank 2252. The realand imaginary part of a single point both use the same address. The fourseparate addresses allow different sets of points to be fetchedsimultaneously during various stages of the transform operation.

As further shown in FIG. 74, the data fetched from conversion RAM bank2252 includes real-and-imaginary pairs denoted X3, X2, X1, and X0. Thedata point pairs being denoted are X3 which includes X3R and X3I, X2which includes X2R and X2I, X1 which includes X1R and XII, and XO whichincludes XOR and X0I. In one embodiment, the data fetched fromconversion RAM bank 2252 is held for two successive multiplier clockcycles in holding latches 2612, so that the same value gets multipliedin each of two multiply cycles. The corresponding twiddle factors arefetched on every clock cycle from twiddle factor look-up table 2610. Forexample in one embodiment, on even clock cycles the real components forfour twiddle factors are fetched in parallel denoted WOR, W1R, W2R, andW3R. These four real twiddle factors are fed to multipliers 2620 and2621,2622 and 2623,2624 and 2625, and 2626 and 2627, respectively asshown, and multiplied by the real and imaginary components of all fourdata value pairs fetched from conversion RAM bank 2252. The products ofthese eight multipliers 2620-2627 are fed through routing logic 2634into all thirty-two adder/subtractor accumulators 2633 of row-columnarray 2632. On the successive odd multiplier clock cycle, four imaginarytwiddle factors are then fetched, shown as WOI, W1I, W2I, and W3I. Asshown, imaginary twiddle factor WOI is fed to both multiplier 2620 andmultiplier 2621. Similarly, imaginary twiddle factor W1I is fed to bothmultiplier 2622 and multiplier 2623. Imaginary twiddle factor W2I is fedto multiplier 2624 and 2625 and imaginary twiddle factor W3I is fed tomultiplier 2626 and 2627. For the second multiplier clock, these eightmultipliers are again fed with the same complex point data still held inholding latches 2612 as was used in the first clock multiplier cycle.Again after the second clock multiplier, the eight products are fedthrough routing logic 2634 and accumulated in adder/subtractoraccumulators 2633. Note that each twiddle factor value in twiddle factorlookup table 2610 is a complex number having a real and imaginary part(in this embodiment, the real and imaginary parts are stored insuccessive locations which are fetched on successive clock cycles, thuspresenting 4 real values followed by four imaginary values) and eachpoint value in conversion RAM bank 2252 is also a complex number havinga real part and an imaginary part (in this embodiment, the real andimaginary parts are stored in side-by-side locations which are fetchedon only every other clock cycle, thus presenting 4 real values and fourimaginary values on every other clock cycle).

These elements are configured to perform a radix-4 butterflycalculation. The radix-4 butterfly calculations needed to provide a1024-point FFT or IFFT and the selection of addresses and twiddlecoefficients for the, for example, 1024 points used for each of the fivepasses needed for a complete 1024-point transform are easily calculatedand are well known in the art (see, for example, chapters 6 & 10 ofTheory and Application of Digital Signal Processing, by Lawrence R.Rabiner and Bernard Gold, Prentice-Hall Inc, published in 1975; and page612 et seq. of Discrete-Time Signal Processing, by Alan V. Oppenheim andRonald W. Schafer, Prentice-Hall Inc, published in 1989). Although aradix-2 transform is also required, a separate radix-2 calculator is notrequired because the radix-2 operation is a subset of the radix-4operation. Only the operands actually needed for the radix-2 operationare actually loaded. The remainder are set to zero. All arithmetic isperformed in such a manner that, if an overflow or underflow shouldoccur, then the results saturate. This prevents roll-over fromcontaminating the results. The function performed by the dual-radix core2600 for a radix-4 butterfly operation supporting the forward transformis the following, where A, B, C and D are the four input points, A′, B′,C′ and D′ are the four output points, and W^(1k), W^(2k) and W^(3k) arethe twiddle coefficients:A′=A+BW ^(1k) +CW ^(2k) +DW ^(3k)B′=A−jBW ^(1k) −CW ^(2k) +jDW ^(3k)C′=A−jBW ^(1k) −CW ^(2k) +jDW ^(3k)D′=A+jBW ^(1k) −CW ^(2k) +jDW ^(3k)

The function performed in a reverse transform is simply the complexconjugate of the above set of equations.A′=A+BW ^(1k) +CW ^(2k) +DW ^(3k)B′=A−jBW ^(1k) −CW ^(2k) +jDW ^(3k)C′=A−BW ^(1k) −CW ^(2k) +DW ^(3k)D′=A+jBW ^(1k) −CW ^(2k) +jDW ^(3k)

Note that the twiddle-factor W which is used for each A on the rightside of the above equations is one (that is, the complex number 1+j0).

The multiplication of two complex numbers, each having a real part andan imaginary part, for example, results in the following equation:X0×WO=(X0R+jX0I)×(W0R+jW0I)=(X0R×W0R−X0I×W0I)+j(X0I×W0R+X0R×WI)+X0R×W0R+jX0I×W0R(for example, the first multiplier cycle in multipliers 2620-2621)X0I×W0I+jX0R×W0I(for example, the first multiplier cycle in multipliers 2620-2621).Thus, four multiplier operations are needed for each complex multiplyoperation.

In order to speed the transform function, the factors for the parallelmultipliers are fetched in parallel under the control of control andclocking sequencer 2640. Routing logic 2634 routes the products of themultipliers 2620-2627 to the thirty-two adder-subtractor-accumulators2633. In one embodiment, eight multiplier cycles, CO through C7, areused to generate four radix-4 butterfly operations, resulting in sixteencomplex output values. For discussion purposes, row-column array 2632 isshown having four rows (A, B, C, and D) and four columns (W, X, Y, andZ) of complex value pairs. The real value and the imaginary value ofeach of these sixteen complex value pairs has its own associatedadder-subtractor-accumulator 2633, for a total of thirty-twoadder-subtractor-accumulators 2633, as shown in FIG. 74 and FIG. 91. Inone embodiment, scaling-factor shift logic 2644 (under the control ofbit-growth selector 2643) is provided in the path betweenadder-subtractor-accumulators 2633 and conversion RAM bank 2252. Thescaling-factor shift logic 2644 provides a right-shift function of 0bits, 1 bit, 2 bits or three bits (divide by 1, two, four or eightrespectively) on each output data value as it is being returned toconversion RAM 2252. Bit-growth pins 2117 through 2118, which controlthe divide function for each of the passes are coupled to bit-growthselector 2643 under control of sequencer 2640.

FIGS. 75-82 are a table 2800 showing the order of calculations for a“normal butterfly sub-operation.” The data points in conversion RAM 2252are arranged within conversion RAM 2252 such that the four input pointsfor one radix-4 operation are each located in different sub-banks if thepoints are successively addressed (for example, addresses 0, 1, 2, and 3are each in different sub-banks, for example, sub-banks 2290, 2291,2292, and 2293 respectively), but points whose addresses differ by afactor of 4 are located in the same bank (for example, addresses 0, 4,8, and 12 are all within bank 2290, as are addresses 0, 16, 32, and 48,addresses 0, 64, 128, and 192, and addresses 0, 256, 512, and 768). Thebutterfly passes for this second set of points (those whoseaddresses-mod-4 are equal) are handled by the equations shown in thetable of FIGS. 75-82. FIG. 75 shows the operations at each of thethirty-two adder-subtractor-accumulators 2633 at a multiplier clockcycle command denoted CO. For example, at CO, theadder-subtractor-accumulator 2633 for the real portion of the AW pointin row-column array 2632 (called the AWR accumulator) gets loaded withthe output (called WR) of multiplier 2620, and theadder-subtractor-accumulator 2633 for the imaginary portion of the DZpoint in row-column array 2632 (called the DZI accumulator) gets loadedwith the output (called ZI) of multiplier 2627. By performing loadoperations at clock CO, the previous values of the accumulators do notneed to be zeroed. As shown in FIG. 91, multipliers 2620, 2621, 2622,2623, 2624, 2625, 2626 and 2627 produce products called WR, WI, XR, XI,YR, YI, ZR and ZI, respectively, however the −R and −I designations ofthese products are not strictly correlated to real and imaginarynumbers. FIG. 91 also shows the row and column locations for thethirty-two adder-subtractor-accumulators 2633, with AWR shown in theupper-left comer and DZI in the lower right corner.

FIG. 76 shows the operations at each of the thirty-twoadder-subtractor-accumulators 2633 at a multiplier clock cycle commanddenoted C1. For example, at C1, the adder-subtractor-accumulator 2633for the real portion of the AW point in row-column array 2632 (calledthe AWR accumulator) gets loaded with the difference of subtracting fromits previous value (called AWR, this value happens to be the WR valueloaded in cycle CO) the output (called WI) of multiplier 2621, and theadder-subtractor-accumulator 2633 for the imaginary portion of the DZIpoint in row-column array 2632 (called the DZI accumulator) gets loadedwith the sum of its previous value (called DZI, this value happens to bethe ZI value loaded in cycle CO) and output (called ZR) of multiplier2626. Similarly, FIGS. 77 through 82 show the operations which takeplace at multiplier clocks C2 through C7, respectively.

Since each complex-multiply operation takes a total of four multiplieroperations, and two multipliers (for example, the pair 2620 and 2621)are used, two multiplier cycles are needed for each complex-multiplyoperation. In a 1024-point transform (that is, either an FFT or anIFFT), four of the five passes involve sets of four points wherein allfour points are in a single sub-bank (for example, 2290), and thereforemust be fetched on four successive even-clocks. Each of these fourpasses takes eight clocks, called C0 through C7. These four passes areeach called “normal butterfly.” Table 2800 shows the order ofcalculation for all of the sub-operations for one embodiment of a normalbutterfly (calculating four radix-4 butterfly operations in eightmultiplier clock cycles), where each of the four points for one radix-4butterfly are in the same sub-bank (for example, either sub-bank 2290 or2291 or 2292 or 2293).

FIGS. 83-90 are a table 2810 showing the order of calculations for a“transposed butterfly sub-operation.” The transposed butterflysub-operation is used for one pass of each transform performed. The datapoints in conversion RAM 2252 are arranged within conversion RAM 2252such that the four input points for one radix-4 operation are eachlocated in different sub-banks if the points are successively addressed(for example, addresses 0, 1, 2, and 3 are each in different sub-banks;sub-banks 2290, 2291,2292, and 2293 respectively). The transposedbutterfly passes for this one set of points (those whose addresses-mod-4are equal) are handled by the equations shown in the table of FIGS.83-90. FIG. 83 shows the operations at each of the thirty-twoadder-subtractor-accumulators 2633 at a multiplier clock cycle commanddenoted CO (note that only eight adder-subtractor-accumulators 2633 areaffected, the other twenty-four do nothing). For example, at CO, theadder-subtractor-accumulator 2633 for the real portion of the AW pointin row-column array 2632 (called the AWR accumulator) gets loaded withthe result of a four-way addition of the outputs (called WR+XR+YR+ZR,these are the realtimes-real portions) of multipliers 2620, 2622, 2624and 2626, and the adder-subtractoraccumulator 2633 for the imaginaryportion of the AZ point in row-column array 2632 (called the AZIaccumulator) gets loaded with the sum/difference of outputs (calledWI−XR−YI+ZR) of multipliers 2621, 2622, 2625, and 2626, respectively. Byperforming load operations at clock CO with no accumulation of the priorvalue (for example, in AWR), the previous values of the accumulators donot need to be zeroed. Note that, since all four points for a singlebutterfly operation can be fetched simultaneously from conversion RAM2252, and the results of the respective multiply operations must all becombined as they are formed, five-way mixed add/subtract operations areprovided for by each adder-subtractor-accumulator 2633.

FIG. 84 shows the operations at each of the thirty-twoadder-subtractor-accumulators 2633 at a multiplier clock cycle commanddenoted C1. For example, at C1, the adder-subtractor-accumulator 2633for the real portion of the AW point in row-column array 2632 (calledthe AWR accumulator) gets loaded with the result of a five-waysubtraction/addition of the outputs (called AWR−(WI+XI+YI+ZI), these arethe imaginarytimes-imaginary portions) of multipliers 2621, 2623, 2625and 2627 and the prior contents of AWR. The adder-subtractor-accumulator2633 for the imaginary portion of the AZ point in row-column array 2632(called the AZI accumulator) gets loaded with the sum/difference ofoutputs (called AZI+(WR+XI−YR−ZI)) of multipliers 2620, 2623, 2624, and2627, respectively.

Similarly, FIGS. 85 through 90 show the four-way and five-way operationswhich take place at multiplier clocks C2 through C7, respectively. Table2810 shows the order of calculation for one embodiment of a transposedbutterfly (calculating four radix-4 butterfly operations in eightmultiplier clock cycles), where each of the four points for one radix-4butterfly are each in different sub-banks (for example, one point insub-bank 2290, one point in 2291, one point in 2292, and one point in2293).

Twiddle-Factor Lookup Table 2610

In one embodiment, the twiddle-factor lookup table 2610 (also called asine-cosine ROM lookup table) comprises 512 fifteen-bit words, whereinfour words can be fetched in parallel. Each complex twiddle factor valuepair is fetched sequentially, wherein the first 15-bit word representsthe real part of the twiddle factor value pair and the second 15-bitword represents the imaginary part, and four values are fetchedsimultaneously (that is, four real values, having 60 bits total, arefetched on an even clock (for example, clocks CO, C2, C4 or C6) and fourimaginary values are fetched on the following odd clock (for example,clocks C1, C3, C5 or C7). In another embodiment, the twiddle factorlookup table 2610 comprises of 256 thirty-bit words. The upper 15 bitsrepresent the real part of the twiddle factor whereas the lower 15 bitsrepresent the imaginary part. Although 1024 complex-value pairs arerequired in order to produce a 1024-point FFT or IFFT, the values arenot unique, and the number of twiddle factors was reduced by a factor offour by making use of the simple trigonometric identities in mapping 360degrees of twiddle factors to a 90-degree lookup table. In oneembodiment, the twiddle-factor lookup table was designed to minimize DCoffset caused by integer-based twiddle factors.

FIG. 92 shows a more-detailed block diagram of anadder-subtractor-accumulator 2633. In one embodiment, multipliers 2620through 2627 are each a 16-bit-by-16-bit multiplier. In one embodiment,only the upper-order 16 bits of the resultant product are passed by MUX2834. (In one embodiment, MUX 2834 is part of router logic 2634.)Addersubtractor 2833 performs a five-way addition/subtraction as definedin FIGS. 83-90 and the two-way addition/subtraction as defined in FIGS.75-82, under the control of sequencer 2640. In one embodiment,accumulator 2835 maintains enough bits above the binary point toaccommodate overflow bits and to provide an indication of overflow whichdoes not get lost as further addition/subtractions are performed on theaccumulating data (in other embodiments, one, two, or three such bitsare maintained).

Input and Output Timing

Tables 7 and 8 below are the detailed timings for one embodiment of theinput and output RAMS.

TABLE 7 Read Cycle for Output RAM bank 2253 Symbol Parameters ConditionNom Tcc Clock Cycle Time Minimum 20 ns Pulse Width Tchpw MinimumPositive Minimum 6 ns CK Pulse Width Tclpw Minimum Negative Minimum 6 nsCK Pulse Width Tavch Address valid to CK Minimum 4 ns high Tchax CK highto address Minimum 1.0 ns change Tchdox CK high to Data Minimum 2 nsOutput change Tcd CK high to data Maximum 15 ns valid Toe Output Enabletime Minimum 0 ns Toz Output Disable time Maximum 7 ns NOTE: The RAMclocks for the input and output banks are limited to 10.24 MHz

TABLE 8 Write Cycle for Input RAM bank 2251 Symbol Parameters ConditionNom T cc Clock Cycle time Minimum 20 ns  Tchpw Minimum positive Minimum6 ns CK pulse width Tclpw Minimum negative Minimum 6 ns CK pulse widthTavch Minimum negative Minimum 4 ns CK pulse width Tchax Address validto CK Minimum 1 ns low Twch CK low to address Minimum 4 ns change TchwWE low to CK low Minimum 1 ns Tdivch CK low to WE high Minimum 4 nsTchdix Data Input valid to Minimum 1 ns CK low Tchdov CK low Data inputMaximum 15 ns  change Tchdox CK low to Data Minimum 2 ns Output changeNOTE: The RAM clocks for the input and output banks are limited to 10.24MHz

Package for ASIC 2101

Package Dimensions and pin out for one embodiment: The ASIC 2101generates 5V TTL output levels and accepts 5V CMOS or 5V TTL inputlevels. TTL Input Levels are defined as follows:VIL max=0.8 VoltsVIH min=2.0 VoltsCMOS Input Levels are defined as follows:VIL max=0.2*VDD(0.9V<VIL<1.1V over 4.5V to 5.5V VDD range)VIH min=0.7*VDD(3.15V<VIH<3.85V over 4.5V to 5.5V VDD range)TTL Output Levels:VOL max=0.4 VoltsVOH min=2.4 Volts

Functional Tests

Testing is broken down into a functional segment to verify devicefunctionality and a scan segment to test for faults in the physicalsilicon. In one embodiment, the vectors are included in a test benchcompatible with LSI Logic's tools. The functions to be tested in thedevice are listed below.

-   -   a. FFT/IFFT Operation—Vectors are provided which are        characteristic of the expected use in the system. The frequency        domain vectors are passed through an inverse transform (with        appropriate bit scaling) and the results stored. The vectors are        then passed through a forward transform (with appropriate        scaling), and this final result analyzed. These tests are        performed for the 1024-, 512-, and 256-point transforms. There        are 10 frames of data for each test. The test bench includes        1024 vectors for the 1024-point transform, 5120 for the        512-point transform, and 2560 for the 256-point transform. There        are twice this number of vectors passed through the device to        complete the test. The total number of test vectors for this        test segment are about 36,000.    -   b. FFT/IFFT Verification—A single sinusoid is passed through the        128-, 64-, and 32-point transforms. Both forward and reverse        directions are tested.    -   c. Bit Growth Tests. Each bit-growth pin (2117-2118) is        exercised for the 1024- and 512-point transform in the forward        and reverse direction.    -   d. Power Down Tests—The device is placed in the middle of a        transform, then powered down. The outputs are evaluated for        correct state. The device is then asked to perform a forward and        reverse transform to validate that the device can function after        the reset.    -   e. Overflow Tests—An overflow condition is induced, and the        device evaluated for correct response (e.g. the overflow pin is        actuated and the event does not cause an adder to wrap around).        The test includes an overflow in the positive and negative        direction.    -   f. Reset—The device is placed into the middle of a transform        operation, then reset. The outputs are evaluated for correct        state. The device is then asked to perform a forward and reverse        transform to validate that the device can function after the        reset.

FIG. 93 is a high-level block diagram of one embodiment of modemreceiver 2402 as shown in FIG. 71. The analog received signal-in isfirst processed by bandpass- and down-convert block 2750. In oneembodiment, the analog received signal-in is either 425 to 600 MHz or550 to 770 MHz, and is converted by bandpass-and-down-convert block 2750to a signal which is 100 kHz on both sides of a 18.432 MHz centerfrequency. In one embodiment, the tuning step size is 99 MHz. In oneembodiment, analog-to-digital decimator system (ADDS) 2850 (in oneembodiment, this is a Sigma-Delta decimator system that uses aSigma-Delta analog-to-digital converter 2840) converts this band-limitedsignal into decimated I and Q quadrature signals, each 15 bits wide,which have a symbol rate of 288 K symbols per second (denoted 288 KBS),which are then processed by FFT block 2849. In one embodiment, FFT block2849 is equivalent to FFT system 2100 of FIG. 70. The outputs of FFTblock 2849 are then processed by post-processing block 2990 into digitaldata out.

FIG. 94 is a more detailed block diagram of modem receiver 2402. Analogreceived signal in is fed to band-pass filter (BPF) 2740 which limitsthe input signal to either 425 to 600 MHz or 550 to 770 MHz. The signalis then amplified by variable-gain amplifier 2741, and mixed by mixer2742 with a demodulator signal of either 627-802 MHz or 752-973 MHzgenerated by signal generator 2747 as controlled by phase-locked-loopblock 2746. The mixer 2742 produces a difference-frequency signalcentered at 202.752 MHz which is filtered by BPF 2743. The resultantsignal is again amplified, this time by variable-gain amplifier 2744,and then mixed by mixer 2745 with a demodulator signal of either 221.184MHz generated by signal generator 2749 as controlled byphase-locked-loop block 2748. The resultant signal is an analog signalcentered at 18.432 MHz and having a 200 kHz bandwidth. The resultant18.432 MHz analog signal is then passed to an analog-to-digitaldecimator which, in this embodiment, is denoted as the componentsencircled by the dotted line referenced as Sigma-Delta decimator system(SDDS) 2850′. The exact mix of components which are included in SDDS2850 can vary from embodiment to embodiment (that is, where the dottedline for SDDS 2850′ is drawn may vary).

In FIG. 94, the 18.432 MHz analog signal is passed through BPF 2839which is centered at 18.432 MHz. The 221.183 MHz signal is divided bydivider network 2838 to produce a 73.728 MHz signal which drivesSigma-Delta converter 2840, and two 18.432 MHz signals (one of which is90 degrees shifted from the other) which drive digital I/Q detector2841. Sigma-Delta converter 2840 is any conventional Sigma-Deltaconverter, such as described in An overview of Sigma-Delta Converters,by Pervez M. Aziz et al., Vol. 13, No. 1, IEEE Signal ProcessingMagazine, January 1996, which is hereby incorporated by reference. SigmaDelta converter 2840 achieves high resolution by oversampling the inputsignal at a frequency much above the Nyquist frequency, and by providinga negative feedback path which uses an analog representation, y_(a)[n]of the quantized output signal y[n]. FIG. 95 shows one such Sigma Deltaconverter 2840, having difference block 2860 that forms u[n] which isinput x[n] minus feedback ya[n]. Signal u[n] then is processed bydiscrete-time integrator 2863 and quantizer 2864, which is simply acomparator. By providing a sampling frequency which is high enough, theSigma-Delta converter 2840 allows the use of a 1-bit quantizer toachieve high overall resolution.

Referring back to FIG. 94, the 73.728 MHz quantized output ofSigma-Delta converter 2840 is coupled to digital I/Q block 2841. In oneembodiment, digital I/O block 2841 is simply two 2-input AND-gates; oneinput of both AND gates is connected to the 73.728 MHz quantized outputof Sigma-Delta converter 2840, the other input of the first and thesecond AND-gate is coupled to the 00 and the 900 18.432 MHz outputs ofdivider 2838. The outputs of digital I/Q detector 2841 are thus twoserial streams, that represent I and Q quadrature signals respectively.The serial I and Q signals are then fed to coarse decimator and MUX2842, which converts the two serial streams into a single N-bit-widetime-multiplexed I/Q parallel stream. In one embodiment, thistime-multiplexed I/Q parallel stream is 10 bits wide. Thistime-multiplexed I/Q parallel stream is clocked at 2.304 MHz, andprovides 10 bits of I followed by 10 bits of Q, thus having 1.152million samples of! interleaved in time with 1.152 million samples of Q.This time-multiplexed I/Q parallel stream is then fed to I/Q demux 2843which de-multiplexes the time-multiplexed I/Q parallel stream intoseparate I and Q streams clocked at 1.152 MHz each, and each 10 bitswide. These separate I and Q streams are then processed by threedigital-processing blocks: DC-offset adjust block 2844 that digitallyadjusts for DC (direct current) balance, DS gain adjust block 2845 thatdigitally adjusts the decimated signal gain, and DS mix block 2846 thatdigitally adjusts the phase.

The Sigma Delta decimator system (SDDS) 2850 is a N-bit A-to-D converterwhich generates a one-bit serial data stream having resolution andaccuracy of N bits (in one embodiment, 15-bit resolution is obtained; inanother embodiment, the AID has a 10-bit resolution with 9-bitlinearity). SDDS 2850 is running on the clock generator 2749 divided to73.728 MHz which oversamples the SDDS input signal in order that it onlypasses data at the 18.432 MHz±100 kHz, approximately. The followingcircuits 2841-2847 then take that 200 kHz of frequency that Sigma-Deltaconverter 2840 passes and shift it down a base band, providing a rangefrom 0-200 kHz. SDDS 2850 turns this relatively slow signal into 10-bitparallel words. The Sigma-Delta converter 2840 outputs a I-bit serialstream, which is ANDed with two 18.432 MHz square waves to produceserial digital I and Q that are two 18.432 MHz gated square waves.

In one embodiment, the entire SDDS 2850 is integrated on a single VLSIchip using 0.8 micron BiCMOS 4S+ technology fabricated by IBMCorporation, with the analog circuits operating from a 5 volt supplyvoltage and the digital circuits operating from a 3.3 volt supplyvoltage. This single-chip implementation facilitates bit growth fromstage-to-stage in order to prevent or reduce truncation or round-offerrors. Thus 10 bit I and Q signals at the output of I/Q demux block2843 are allowed, in one embodiment, to grow to 25-bits at the output ofDS mix block 2846 through the digital processing of DC-offset adjustblock 2844, DS gain adjust block 2845, and DS mix block 2846. Forexample, the N bits each of the I and Q data streams at the output ofI/Q demux block 2843 grow a little to N+ bits after the digitalprocessing of DC-offset adjust block 2844, N″ bits after DS gain adjustblock 2845, and N″′ bits after DS mix block 2846. Decimators 2847 and2847′ select one out of every four values from the I and Q data streams,respectively, thus producing a 288 kHz rate of 25 bits each for the Iand Q streams. These then pass through scaling blocks 2848 and 2848′which scale each data stream to 15 bits, which are denoted 1′ and Q′ andare coupled to 15-bit FFT 2849.

One consideration with the large number of signals on such a single-chipimplementation is to minimize the number of different clock signals. Inthe embodiments described for FIG. 94, for example, this is accomplishedby running a large number of blocks from a single clock, even thoughsome of those blocks might be able to run off a different and slowerclock.

Overview of Data-Delivery Architecture

Referring now to FIGS. 96 and 97, there is shown the overallarchitecture of a data delivery system 500 according to anotherembodiment of the invention. The data delivery system 500 provides highspeed data access to the home 510 or office 512 (alternately referred toabove in FIG. 1 as remote units 46) over the HFC distribution network 11using the infrastructure of the modem-based telephony transport system10 described hereinabove. FIG. 97 illustrates the integration of thedata delivery elements in the HDT 12. The system allows users to haveaccess to local content 520 and the Internet 530 through servicesavailable at the cable providers' premises or head end 32.

Among other things, system 500 provides: (1) user data access to theInternet 530 and local content on a head-end server 520 through theabove-described access platform; (2) support for TCP/IP and transparentbridging at the data link layer using a Dynamically Adaptive TransportSwitching (DATS) methodology (described below); (3) guaranteed, reliablesymmetrical data service at transfer rates from 64 Kbps to 512 Kbps, inincrements of 64 Kbps, for geographically dispersed individuals; (4)guaranteed, reliable, symmetric shared access to a 8.192 Mbps data pipefor geographically limited group of users connected in a routedconfiguration; (5) mixing of data and telephony within a single HDT 12;(6) network management for telephony, video, and data through integratedCMISE and SNMP; (7) routed service through a head end server; and (8)use of HISU and MISU RF modem technology for transport.

Referring now to FIGS. 98-100, the Cable Data Modem (CDM) 535 for thesystem 500 preferably can take three forms: a stand-alone box called thePersonal Cable Data Modem (PCDM) 540, a HISU add-in card called the DataModem Service Module (DMSM) 550, or a MISU add-in card called the DataModem Channel Unit (DMCU) 560. The stand-alone PCDM 540 has severalconnector options; including standard F style coax cable, 10baseTEthernet 542, and RS232 544. The DMSM add-in card preferably resides inan expansion slot on HISU 68 enclosure and will terminate a customer'scomputer with a 10BaseT Ethernet connection. The DMCU 560 add-in cardwill reside in a line-card slot of the MISU and will provide a 10BaseTEthernet routed connection to four users. Customers without anintegrated Ethernet port can add one to their system at minimal cost.

The system 500 provides connection to head end 32 services via the DATSmethodology. The DATS methodology intelligently allocates bandwidth inthe system to maximize both transport bandwidth and number of userswhile providing guaranteed bandwidth. Under TR303N5 switchingenvironments a central resource within the HDT 12 (described below)provides the intelligence to allocate bandwidth and efficiently managetransport load. This capability is implemented at the customer end byinitiating a connection to the head end 32 when data is available tosend. When the session is initiated, the head end 32 equipmentdetermines the amount of bandwidth to be allocated to the subscriber asconfigured during pre-provisioning. The connection is maintained asneeded and dropped when transmission is complete. During the time thatthe connection is maintained, bandwidth is guaranteed, providing theefficiency of packet switching and the guaranteed bandwidth ofconnection switching. All processing is performed through standardTR303N5 call processing and therefore integrates completely withtelephony provided over system 500. The architecture of the system 500provides guaranteed bandwidth and latency to all supported users. Asindicated in FIG. 101, up to 24 concurrent users can be supported at 512Kbps within a single 6 MHz transport channel.

Using the DATS methodology, bandwidth is dynamically allocated tosupport a maximum of 240 users per 6 MHz channel (for example, see FIG.13) at 64 Kbps. Each user on the system is guaranteed the allocatedbandwidth for the duration of the session. As such, the bandwidth andlatency seen by the user remains constant, independent of the trafficload, application type, or number of users. Assuming even distributionacross all applications, the average frame size carried downstream is378 bytes, while the average upstream frame size is 69 bytes.

The downstream latency from the HDT 12 is dominated by filling a 512 Kdata pipe, all other latencies in the system are insignificant.Therefore on average the latency to transfer a TCP/IP frame from the HDTto the customer is 5.9 ms. With Forward Error Correction (FEC) disabled,the average latency in the upstream direction, from the customer end tothe HDT is also dominated by filling a 512 K data pipe. Therefore, onaverage the latency to transfer a TCP/IP frame from the HDT to thecustomer is 1 ms. Adding FEC will add up to 7 ms to the upstreamlatency.

Data Transport and Framing

The system 500 provides transparent bridging and data transport. Aschematic representation of the data transport and framing is shown inFIG. 102. Upstream data traffic begins at the customer end equipment 511where frames are put out on the Ethernet link 542. The CDM 535 buffersthe frames and encapsulates them with HDLC framing 570. The HDLC frame570 is then sent out over the coax by a modem 101 as one or more of thepossible 240 payloads on the coax. The frame, along with other data andtelephony streams is converted at the ODN 18 and sent to the head end 32over optical fiber. At the head end, the CXMU 56 demodulates and sendsthe stream to the CTSU 54, where it is routed to the appropriate LANU580 as part of a MARIO data stream. The LANU 580 buffers the completeframe, stripping the HDLC framing prior to putting the frame on thelocal Ethernet. All routing for the Internet is performed by thehead-end server 590. The routing services could also be incorporated inLANU 580.

In the case of downstream traffic, the head-end server 590 puts Ethernetpackets on the LAN. One to potentially seven LANUs 580 examine theEthernet address and selectively pick up packets destined for thecustomer-end machines. Once the Ethernet packet has been accepted, thepacket is framed in HDLC and sent, via a MARIO stream, to the CTSU 54according to the routing established when the call was initiallyprovisioned (explained further below). The CTSU 54 forwards the streamto the CXMU 56 where it is sent over the HFC distribution network 11 tothe customer premises equipment 511. At the CDM 535, all packetsreceived are passed on to the local Ethernet connection 542. In the caseof a broadcast message, the LANU 580 at the head end will replicate thepacket across all attached connections. This allows the bridge tosupport protocols such as ARP.

The system 500 will utilize the telephony error correction mechanismdescribed above with respect to system 10 of FIG. 1. Under the telephonyerror correction scheme, forward error correction codes are generatedfor upstream traffic but not for downstream traffic. Forward errorcorrection on upstream is generated at the ISU 100 (HISU 68 or MISU 66)and consumes the 10th bit of each DSO, thereby protecting each DSOseparately. The error correction can be disabled, but this is notrecommended for data transport.

The error detection/correction processing occurs on the CXMU 56 of FIG.3 and corrected data is delivered to LANUs 580, DSIUs 48, in MARIOstreams. Therefore, the system 500 data architecture does not explicitlyhave to deal with error correction. The CRC of the HDLC frames providefor a level of error detection above the error detection/correction ofthe CXMU 56. Errors detected in the LANU 580 will be reported throughthe SNMP agent of the LANU 580.

Component Design

The data delivery hardware for the head end 32 equipment consists of theLANU 580 which interfaces with the HDT 12. The LANU 580 includes a boardresponsible for all data transport to and from the head-end server LAN591. In that function, the LANU 580 operates as a point-to-multipointconnection that is responsible for concentrating up to 128 DSOs onto asingle Ethernet connection. The LANU 580 has the following features:

-   -   (1) It sits in a DS1U 48 slot and is form/fit compatible with        the HDT 12 backplane.    -   (2) It provides 10BaseT Ethernet connection to head-end LAN.    -   (3) It supports multiple LANUs 580 on a single LAN through a        10BaseT connection or fast Ethernet through hub.    -   (4) It concentrates up to 128 DSOs into a single Ethernet        Connection.    -   (5) It supports time slot assignment and super-channel        aggregation across four MARIO data streams independent of the        CTSU 54.    -   (6) It supports Dual HDT 12 LAN ports.    -   (7) It maintains on-board FLASH for storage of operational code        image.    -   (8) It supports Nx64 service for super-channels up to 512 Kbps.    -   (9) It guarantees time ordering of multi-channel calls        independent of transport system.    -   (10) It provides transparent bridging and broadcast of Ethernet        frames between head-end LAN 591 and CDM 535.    -   (11) It self-discovers Medium Access Control (MAC) addresses of        CDM 535 and filters Ethernet frames with on-board CAM.

A high-level block diagram of the LANU 580 is shown in FIG. 103. TheLANU 580 preferably employs a single processor, unified memory design.In this embodiment, the main processor on the LANU 580 is the MotorolaMC68 MH360 QUICC32 (581), running at 33 MHz, which can deliverapproximately 5 MIPS. The QUICC32 is actually a dual processor embeddedcontroller that also contains a dedicated independent RISC processorcommunications called the CPM. The CPM along with Direct Memory Access(DMA) can access memory and move communication traffic without placingany performance burden on the main processor. In addition to acting asthe host processor, the QUICC32 can also act as a slave. The LANU 580implementation is preferably based on 4 QUICC32s with one acting asmaster and the remaining three operating in slave mode. The combinationof the four QUICC32s provide the majority of the system peripherals.

There are several types of memory 582 on the LANU 580. The first andlargest is 8 MB of page-mode DRAM for the storage of LANU operationalcode. In addition to operational code, the DRAM also holds the routingtables, data buffers, and buffer descriptors needed to process datatraffic. Second, the LANU contains 2 MB of FLASH memory to store animage of its operational code. On power-up, the image is loaded intoDRAM. Finally, the LANU also contains 128 Kbytes of EPROM to hold bootcode. The boot code will load the contents of the FLASH into DRAM onpower-up. If greater performance is desired, fast SRAM can be added tohold the data buffers and buffer descriptors for the data traffic.

There are two sets of system peripherals on the LANU: those that arecontained within the QUICC32s and those that are not. Regardless oflocation, peripheral control on the LANU 580 is managed by the SystemIntegration Module (software) (SIM) of the master QUICC32. The SIM isresponsible for managing generation of Chip Select (CS) and othercontrol lines.

The most important of the system peripherals are the QMC HDLCcontrollers 586 located in the QUICC32s. The QMC of the QUICC32 canimplement up to 32 HDLC controllers running at 64 Kbps and are used toprocess the data traffic in the MARIO streams from the CTSU 54. The QMCs586 can also support super channels by aggregating multiple 64 Kbitchannels into higher data rate channels. The QMC 586 is implemented inthe CPM of each QUICC32 and with the addition of SDMA can autonomouslymove communications data to and from main memory 582. Operation of theQMC 586 is controlled by the master processor 581 through the use ofbuffer descriptors that reside in external memory 582.

In addition to the QMC 586, three additional HDLC controllers 587 areused to provide two 2 Mbps HDT backplane LAN ports and a 10 MbitEthernet port. As with the QMC 586, the master processor 581 controlsthe HDLCs through the use of buffer descriptors, but unlike the QMC,these data structures reside in dual ported RAM (DPR) that are internalto the QUICC32s.

There are two additional system peripherals that reside outside of themaster and salve QUICC32s. The first is the Rate Adaptation/Ninth BitSignaling/Time Slot Assigner logic block (RA/NBS/TSA) 588. TheRA/NBS/TSA 588, as its name suggests has three functions. The rateadaptation function is responsible for converting the 2.56 Mbps, 9 bitdata format of the input MARIOs to the 2.048 Mbps, 8 bit data format ofthe QMC 586 in both the receive and transmit directions. In bothdirections the RA is also responsible for managing any information thatmay be placed in the 9th bit.

The NBS is responsible for transmitting and receiving the data that iscarried with each DSO in the ninth bit. The ninth bit is used for twodistinct functions. In the upstream direction the ninth bit carriesinformation regarding the ordering of data within a multi-channel call.The signaling consists of a repeating number that indicates which timeposition the DSO occupies in the multi-channel call. The format of theupstream ninth bit signaling is shown in Table 9.

TABLE 9 Upstream Ninth Bit Signaling Bit Contents Description 1 “1” SyncPattern 2 “1” Sync Pattern 3 “1” Sync Pattern 4 “1” Sync Pattern 5 “0”Sync Pattern 6 “0” Sync Pattern 7 D[8] Order Number, Bit 8 [MSB] 8 D[8]*Inv. Order Number, Bit 8 [MSB] 9 D[7] Order Number, Bit 7 10 D[7]* Inv.Order Number, Bit 7 11 D[6] Order Number, Bit 6 12 D[6]* Inv. OrderNumber, Bit 6 13 D[5] Order Number, Bit 5 14 D[5]* Inv. Order Number,Bit 5 15 D[4] Order Number, Bit 4 16 D[4]* Inv. Order Number, Bit 4 17D[3] Order Number, Bit 3 18 D[3]* Inv. Order Number, Bit 3 19 D[2] OrderNumber, Bit 2 20 D[2]* Inv. Order Number, Bit 2 21 D[1] Order Number,Bit 1 22 D[l]* Inv. Order Number, Bit 1 23 “0,, Synch Pattern 24 “0,,Synch Pattern

The numbers can range from 1 to 128 with a 0 indicating that the DSO hasnot been assigned a position. During a call, the processor 581 willmonitor the ninth bit signals from all the channels and once the orderhas been established, the processor will configure the TSA function toorder the channels. Once the order has been established, the processorwill periodically monitor the ninth bit signaling to detect any changesin ordering (that is, frequency hoping due to excessive errors). In thedownstream direction, the NBS is used to enable data transmission. Oncethe LAND 580 receives the ordering information for the channels, theprocessor will enable transmission by sending a data pattern over thedownstream ninth bit of the first DSO of a multi-channel call. Theformat for the “Data Dial Tone” signaling is shown in Table 10.

TABLE 10 “Data Dial Tone” Downstream Ninth Bit Signaling Bit ContentsDescription 1 “1,, Sync Pattern 2 “1,, Sync Pattern 3 “1” Sync Pattern 4“1” Sync Pattern 5 “0” Sync Pattern 6 “0” Sync Pattern 7 CMD[15] CommandBit 15, MSB 8 CMD[14] Command Bit 14 9 CMD[13] Command Bit 13 10 CMD[12]Command Bit 12 11 CMD[11] Command Bit 11 12 CMD[10] Command Bit 10 13CMD[9] Command Bit 9 14 CMD[8] Command Bit 8 15 CMD[71] Command Bit 7 16CMD[6] Command Bit 6 17 CMD[5] Command Bit 5 18 CMD[4] Command Bit 4 19CMD[3] Command Bit 3 20 CMD[2] Command Bit 2 21 CMD[1] Command Bit 1 22CMD[0] Command Bit 0, LSB 23 “0” Synch Pattern 24 “0” Synch Pattern

Two commands are defined. The first is “Idle Sync” (bit pattern forCMD[15:0] is 0000 0000 0000 0000) is sent during idle times tosynchronize the receivers. All idle syncs sent from the LAND 580 will besynchronized themselves to simplify the reception of order informationin the upstream ninth bit signaling. The second command is “Data DialTone” (bit pattern for CMD[15:0] is 1000 0000 0000 and is sent once theorder of the super-channel has been established. This command willinstruct the CDMs to enable transmission.

The TSA is responsible for ordering and aggregating channels that aresent to the QMCs 586. Although TSA has the ability to aggregate up to128 channels into a single data stream, most applications will aggregatemultiple super-channels, up to 8 DSOs per super-channel, among fourDS1-like channels feeding the four onboard HDLC controllers. For 1 to 32DSO data streams, the DSOs are aggregated and sent to the QMC 586 as a2.048 Mbps serial stream. Within a single DSI-like stream, multiplesuper-channels can be supported by connecting to a single QUICC32 QMC586. The TSA can also aggregate 128 DSOs into a single 8.192 Mbps datastream and connect it to a QUICC32 HDLC controller. In both cases theTSA is configured to insure that the time ordering of the data ismaintained.

The final peripheral that does not reside in a QUICC32 is the ContentsAddressable Memory (CAM) 589. The CAM performs memory accesses basedupon data rather than address and is used to quickly determine whetheran Ethernet frame should be accepted. The Ethernet controller 587 ainterfaces to the CAM 589 through glue logic and the reject input. Whena frame is received that is not in the CAM, the CAM logic asserts thereject control line and the received portion of the frame is discarded.The buffer depth of the Ethernet controller is set so that no memoryaccesses are generated on rejected frames. The CAM is available off theshelf from MUSIC semiconductor.

The LANU 580 sits in a DS1U 48 slot at the HDT 12 and is form and fitcompatible with the DS1U 48 to be backplane compatible. The LANU 580 hasfour major system connections: MARIO 592, Ethernet 593, HDT LAN 591, andclocking (input not shown in FIG. 103). The four MARIO connectionsconnect the LANU to the CTSU over the backplane. The four MARIOconnections provide up to 128 bidirectional, 64 Kbit channels. TheEthernet connection connects the LANU to a local 10 Mbit, 10BaseT LAN.The 10BaseT connection will take the place of a single T1 connection ofa DS1U. The connection is terminated by wire-wrapping the 10BaseT to thebackplane and routing to a patch panel. For call provisioning and othersystem functions, the LANU 580 connects to two common 2 Mbps HDLC LANson the backplane. Finally, system clocking is provided by an externalsource such as a Building Integrated Timing Supply (BITS).

Functional Description of Upstream Communications

For upstream traffic, the HDLC framed data available on the MARIOinterface passes through the rate adaptation and TSA block 588. In thisblock, the 2.56 Mbps MARIO interface is rate adapted down to 2.048 Mbps.As part of rate adaption, the ninth bit of each DSO+ of the MARIO streamis stripped and sent to the NBS logic. The ninth bit carried an ordernumber that is used to time order the DSOs in multi-channel calls. Oncethe order numbers are established, the processor 581 configures the TSAto re-order the multi-channel calls and target the super-channel to aQMC 586. For super-channels composed of 32 or less DSOs, the call isplaced in a single 2.048 Mbps data stream, along with other calls andsent to a QMC 586. For 128 DSO calls, the DSOs are placed in a single8.192 Mbps stream that is target to a QUICC32 HDLC controller 586configured for 8.192 Mbps HDLC. Whether targeted to a 2.048 Mbps QMC or8.192 Mbps HDLC, the frames are accumulated and transmitted on the localEthernet LAN.

Functional Description of Downstream Communications

For downstream traffic, data on the LAN is filtered according to thedestination MAC address. If the MAC address is in the ContentAddressable Memory (CAM) 589, the LANU 580 will accept the Ethernetframe. Once the frame is accepted, the LANU 580 accesses a routing tablein memory 582 to select the appropriate MARIO slot for transport. Theframe is then scheduled for transmission and the HDLC controller 586takes over. In the downstream direction for 32 DSO or less calls, theHDLC controller 586 is responsible for creating the MARIO stream andencoding the data into HDLC format. For 128 DSO calls, the 8.192 MbpsHDLC data stream is split among the four MARIO interfaces (A-D). Thetime ordering and aggregation in the downstream direction is controlledby the TSA. After the data passes through the TSA, the ninth bitsignaling information is added to indicate that data transmission isenabled. At the same time that the ninth bit signal is added, the datastream is rate adapted up to the 2.56 Mbps rate of the MARIOs.

For data movement the on-board memory 582 provides the switching andbuffering necessary to move traffic from the Ethernet to the MARIO andvice-versa. The data switching provides a virtual circuit based uponrouting information constructed during call provisioning that maps theMAC address to MARIO time slot. The LANU 580 constructs this routingtable from information provided during call provisioning and byexamining the source MAC address of Ethernet frames received from thecustomer end. The MAC addresses supported by the LANU are then placed inthe CAM 589 where they can be used to filter Ethernet traffic.

In addition to the MARIO and Ethernet interfaces, the LANU 580 alsosupports two HDT backplane LAN 591 interfaces at 2.048 Mbps. The HDT LANis maintained by the SCNU 58 at the head end 32 and is used to passconfiguration information to the LANU 580. The combination of-LANU 580,call processing, and pre-provisioning will provide the MARIO route (timeslot in the MARIO data stream) to all customer devices.

Customer Premises Equipment (CPE)

The CPE of the system 500 comes in different implementations. Althoughthe basic data framing and transport remain the same across all theimplementations the underlying modem technology and form factor differs.There are currently three general types of CPE defined: HSU based, MISUbased, and Stand-alone (a variant of the HISU based implementation).Each of these implementations are discussed separately below.

Data Modem Service Module

A simplified block diagram of the DMSM is shown in FIG. 104. DMSM 550supports up to 8 DSO's for data traffic. Data is interfaced to the HISU68 through a Service Growth Module (SGM) interface that implements thebridge router functions of the data connection. The bridge/router isbased upon a 68302 embedded processor 605 with 2 MB of DRAM 604 and 256Kbytes of FLASH memory 606. One embodiment of the customer interface isa 10BaseT, 10 Mbit Ethernet connection 602. The interface between thebridge/router and the HISU modem is a Super-LUIGI (SLUIGI) interfaceoperating at 2.048 Mbps. Data in the upstream direction is accepted bythe Ethernet connection and relayed to the HISU 101 modem over theSLUIGI interface as HDLC encoded data. In the downstream direction, datafrom the SLUIGI interface is passed to the HDLC controller 607 and thensent out the Ethernet connect through Ethernet controller 609. Systemmemory consists of 2 Mbytes of DRAM 604 which contains the operationalcode that implements the bridge/router functions as well as an SNMPagent. The 256 KBytes of FLASH 606 is used for storage of theoperational image and can be updated with TFTP transfers.

In the upstream direction, the HISU interface logic 608 is responsiblefor generating the ninth bit signaling information for each DSO andinterfacing with the SLUIGI stream of the HISU. The 2.048 Mbps serialdata from the HDLC controller is stuffed in the appropriate time slotsof the SLUIGI interface. In the case of 64 Kbit traffic, all HDLC datais placed in a single SLUIGI time slot. In the case of multi-channelscalls, all HDLC is placed in adjacent SLUIGI time slot receiving thefirst byte of HDLC data. In addition, the interface logic will generatean order number for each DSO with “1” assigned to the first DSO and “8”assigned to the last DSO, “0” is reserved to indicate that the positionof the DSO has not been set. In the event of frequency hopping, theinterface logic will continue to number the DSO time-slots in the orderthat they are received. For example, under normal circumstances DSOninth bit signaling would tag the time slots as “12345678-” for an eightchannel call. If DSO 4 is lost due to ingress, then the new order wouldbe “123-45678.” This signaling information is used at the head-end toreorder the DSOs independent of the frequency hopping and transportservices. In the downstream direction the HISU interface logic 608 isresponsible for taking data from the SLUIGI interface from the RF modem101 and giving it to the HDLC controller. In addition, the HISUinterface logic monitors the ninth bit signaling information of thefirst DSO to detect the “Data Dial Tone” sequence. The “Data Dial Tone”sequence is sent by the head-end to enable data transmission. During thecall provisioning process the HISU interface logic sends the ninth bitordering information as soon as the SLUIGI time slots indicate that theycan accept data. It is not until the HISU interface logic gets apositive acknowledgment through the “Data Dial Tone” that data is sentto the head-end in the upstream direction.

Data Modem Channel Unit (DMCU)

A simplified block diagram of a DMCU 610 is shown in FIG. 105. The DMCUsupports up to 128 DSOs for data traffic. The data interface to the MISUis a specialized channel that sits on the MISU backplane. The basicdesign of the DMCU 560 is very similar to the design of DMSM 550. Themost notable difference is the interface to the RF portion is two 8.192MHz serial channels. This allows the MISU interface to support asymmetrical 8.192 Mbps Ethernet connection. Because of the higherthroughput the MISU is based on the MC68360 614 that can support boththe 8.192 Mbps HDLC connection as well as the 10 Mbit Ethernet. In frontof the 10 Mbit Ethernet interface 612 is a router 616 that allows fourusers access to the 8.192 Mbit data connection. The router 616 designensures security for all connected users. The design contains 2 Mbytesof DRAM 618 and 256 bytes of FLASH 619. Like the HISU design, the FLASHcan be remotely updated with TFTP.

The DMCU 610 has an equivalent interface function that moves data fromthe HDLC controller 611 to the RF modem and works in a very similar wayto the DMSM 600 with the exception that the MISU modem interface isformatted as two SLUIGI streams that are clocked at 8.192 Mbps. Betweenthe two SLUIGI streams, 128 DSOs can be carried between the HDLCcontroller 611 and the modem. The MISU interface logic 613 isresponsible for buffering and then sending data over the dual SLUIGIinterface to and from the RF modem. In the upstream direction the MISUinterface logic 613 generates order numbers for each of the 128 DSOsover the ninth bit of the DSO+(the order numbers generated on the MISUwork in the same way as they do on the HISU). In the downstreamdirection, the MISU interface logic 613 is responsible for moving datafrom the S-LUIGI streams to the HDLC controller 611. The MISU interfacelogic also monitors the ninth bit data stream from the first DSO todetect the “Data Dial Tone” that enables data transmission. Ethernetcontroller 617 moves the data to the router 616.

Stand-Alone Data Modems

The stand-alone data CDMs are based upon the HISU design. In thestand-alone designs, the RF modem 101 of the HISU is tightly integratedwith the bride/router design. Like the DMSM 600, the stand-alonesupports from 64 K to 512 K. The interfaces are identical with severaloptions: standard “F” style connector to the cable, 10 Mbit Ethernet andRS232 connection to the customers equipment.

System Software and Call Processing

In the TR-008N2 system, calls are provisioned and nailed up at time ofinstallation. Under this scenario an operator at the head-end 32 isresponsible for determining the MARIO configuration and transfer rates(64 K to 512 K). The DATS methodology of present invention utilizesTR-303N5 call processing to provide dynamic allocation of bandwidth. Tomaintain the telephony oriented architecture of the access platform ofthe present invention, the LANU 580 takes on responsibility of a limitedsubset of the Central Office (CO) functions. This approach has thedistinct advantage that the data sessions are fully integrated withtelephony.

At the time of deployment, a LANU 580 will be identified as a “Master”LANU (mLANU). The mLANU will have the responsibility to maintain theCO-like functionality for all data calls to the HDT 12. In order toperform these functions, the mLANU will represent itself to the HDT asan IDT. When the mLANU is pre-provisioned, the mLANU will be given anIDT identifier and assigned a Time Slot Management Channel (TMC) channelfrom the CTSU 54. Regardless of the number of LANUs 588 in the HDT 12 asingle mLANU will allocate and keep track of available DSOs for all theLANUs in the HDT. As customers are configured, a Call Reference Value(CRV) for the selected CDM will be assigned to identify the customer.The CRY along with the number of data channels will be added to a callprovisioning database on the mLANU.

The call processing sequence for call origination is shown in FIGS. 106109. Call processing begins when the CDM generates an “Off-Hook” messageover the IOC associated with the HISU, MISU, or Stand alone CDM(described above). After the “Off-Hook” message is received at the CXMU56 then sends a “Request Service” message over the backplane. LAN 591 tothe CTSU 54 identifying the CRY of the originator. After receiving the“Request Service” message, the CTSU 54 sends a set-up message to themLANU 580 over the TMC (DS0 in a MARIO stream). The mLANU uses the CRYto access the on-board database in memory 582 and determine the numberof DSOs to allocate the call. Once the number of channels has beendetermined, the mLANU identifies the DS0 and DS1 for the call. The mLANUthen sends a “Make Cross Connect” message to the CTSU 54 over the TMCidentifying the DS0 and DSI and their association with the CRY. Inresponse the CTSU 54 sends a “Req. Bandwidth” message to the CXMU 56over the backplane LAN 591 to allocate the bandwidth in the transport.

Preferably, the DAT methodology and system provides that each subscriberis represented in the database as having subscribed to a certain levelof bandwidth per data connection. For example, a subscriber may sign upfor 512K of bandwidth. Upon call setup or connection, the DATmethodology thus assigns each user the number of channels required toachieve the subscriber's bandwidth. However, in certain cases, thetransport system 500 may not have the necessary bandwidth to allocate toa subscriber their normal subscripted bandwidth. Under thesecircumstances, the subscriber is allocated a lesser amount of bandwidth,for example 64K of bandwidth. By dynamically adjusting the amount ofbandwidth assigned at call setup for each call, the mLANU can maintain aminimum level of bandwidth for each subscriber. However, existingconstraints prevent small decremental bandwidth re-allocations.Constraints existing within the system 500 as described above willenforce the halving of bandwidth of some subscribers to accommodateadditional subscribers, rather than by means of a more evenlydistributed loading. As shown in FIG. 107, as the number of subscribersincrease (on the horizontal axis) any given user's allocated bandwidthwill halve at a certain user density. The spread around the averageindicates that some users will necessarily lose half their existingbandwidth earlier under loading than will others.

The constraining factor in preventing a more equitable burden is thewindow nature of the present HISU 68 RF tuning. As noted above, the HISUtunes to one of 24 IOC channels spread throughout the 6 MHz cablechannel and has access to five payload channels above and five payloadchannels below the selected IOC frequency. It cannot borrow payloadchannels outside this window of ten channels, so therefore there is noway for the “25th” user to borrow just one channel each from seven otherusers. It can only sit on top of one of the existing IOC payload windowsand takeover half of the window bandwidth. At that point, 23 users wouldbe granted 512 kbs bandwidth and two users would each get 256 kbsbandwidth. The “26th” user would result in 22 users with 512 kbs andfour users with 256 kbs, and so on. This general pattern is repeated ata load of 72 users and 120 users. (There is a discontinuity in thepattern from 48 to 72 users due to the previously unused two often DSO'sper window being pressed into two DSO 128 kbps service.) The graph ofFIG. 109 illustrates the distribution of bandwidth to users as thenumber of users increases. It is also contemplated that a subscribercould have different default or standard data rates depending on thetime of day or day of week, or based on system loading, such that a usercan receive even more bandwidth than their standard rate under certainsystem loading conditions, such as if the system is loaded below apredetermined threshold at the time the subscriber seeks a connection.Also, using the ninth bit signaling, the mLANU can “steal” bandwidthfrom other (e.g. high capacity) users. This is done by removing the DataDial Tone from a subscriber using the ninth bit signaling. This quiescesthe user's line, allowing, the number of channels for that user can bereassigned to increase the bandwidth allocated to the user. Thistechnique can also be used to decrease the number of channels assignedto a user.

In order to establish the transport, the CXMU 56 trains the modems (asdescribed above with respect to system 10) and associates the availabletones with DSOs. Once the training is complete, the CXMU 56 sends a“Pass” message to the CTSU 54, which in turn informs the mLANU over theTMC that the call is complete with the “Call Complete” message. Inresponse, the mLANU configures the HDLC controllers 586 and the TSA 588on the mLANU or another LANU through communications over the backplaneLAN. At this point the pipe is established but data is not yet enabled.

In order to actually begin data transmission two additional steps haveto occur across the ninth bit signaling of the DSOs. At the point wherethe modems are trained the HISU 68 or MISU 66 interface logic (608,613)will be enabled to transmit data. Once the transmit is enabled, theinterface logic will begin transmitting the DSO ordering number in theninth bit of each DSO. At the LANU 580, the processor 581 will monitorthe ninth bit signaling to determine when all DSOs have establishedtheir order. Once all DSOs have established order, the LANU 580 willsend the “Data Dial Tone” pattern on the ninth bit of the first DSO inthe multi-channel call. When the ISU 100 receives the “Data Dial Tone”data communications are enabled and data transmission begins.

A session is terminated at the customer end when no data is availablefor transmission by generating an “On Hook” message. The call processingsequence for an “On Hook” message is shown in FIG. 110. When the CDMterminates the connection, an “On Hook” message is sent over the IOC tothe CXMU 56. The CXMU 56 in response sends an “On Hook” message,identifying the CRY, to the CTSU 54. The CTSU 54 then sends a “TearDown” message to the mLANU 584 over the TMC. At the mLANU 580, theconnection is deleted from the connection database and then released. Ifthe connection is not on the mLANU, the mLANU will send a “ReleaseChannel” message to the target LANU 580 and also will send a “ReleaseCross Connect” message to the CTSU 54. The CTSU 54 will release thecross connects used for the connection and then send a “ReleaseBandwidth” message to the CXMU 56. At the CXMU 56 the mapping betweentones and DSOs is lost and the connection is lost. When the connectionis lost, the CDM will lose the “Data Dial Tone” in the ninth bitsignaling of the first DSO of the call.

The LANU 580 can also be configured to bring up connections tocustomer-end equipment. This allows for notification of incoming e-Mailand personal Web pages without tying up idle bandwidth. To do this, themaster LANU in each system will maintain a mapping between the MACaddress for each data element in the system and cross that with the CRY.Then an Ethernet packet is put on the head-end LAN 591, and a LANU 580will read its MAC address and determine whether the connection is up tothe device. If the connection is up, the packet will be forwarded overthe HFC transport. If the connection is not in place, the receiving LANU580 will generate a connection request to the mLANU. The mLANU will thensignal the transport system over the TMC to bring up a connection tothat device using the IOC. The receiving LANU 580 will then send thedata once the connection has been established.

LANU Software

The LANU software 620 is responsible for the all major function of thedata concentration of the head-end equipment. A simplified schematicdiagram of the LANU software is shown in FIG. 111. The software 620consists of three major components: bridging 621, HDLC LAN manager 622,and data IDT 623. All three tasks will operate as applications on top ofthe embedded controller operating system “pSOS” kernel in the processor581. The pSOS kernel will provide the base for the multi-taskingoperation of the software 620.

The most important task to the actual transport of data will be thebridging task. The bridging task has several functions. First, the taskwill be responsible for providing the virtual switch between the MARIOand Ethernet interfaces. The task will be implemented as an “interrupton receive” task that will execute at interrupt level. At eitherinterface, an interrupt is issued when an entire frame has been receivedand stored in buffer memory (FIG. 103, 582). During the interruptservice routine, the packet will be handed off by modifying theassociated buffer descriptor after looking up the routing in thebridging table (stored in memory 582). For upstream traffic (HDLC toEthernet), the source of the first packet will be read to discover itsMAC address. This address will be added to the bridging table andwritten to the Ethernet CAM 589 for filtering.

A second function of the bridging task is the creation and maintenanceof the bridging table. The bridging table will match the MAC address ofthe CDM 535 with the MARIO DSOs so that data can be moved between theEthernet and HDLC. During call processing, the DSOs that are allocatedto the call will be identified by the mLANU 580 through backplane LAN(591) Messaging and installed in the bridging table. As described above,when the first frames begin to flow from the CDM, the source MAC addresswill be identified and the table entry for the CDM will be complete. Atthis point data will flow in both directions. Once the MAC address hasbeen discovered, the bridging table entry will remain intact until theconnection is terminated by the CDM.

A third function of the bridging task is maintenance of an SNMP agent.SNMP traffic will be handled and processed from the Ethernet port. Theagent will support a standard MB-II information database for transparentbridging. In addition, objects will be added to the MIB that arespecific to the data architecture to facilitate CMISE-SNMP integrationand different billing options.

Finally, the bridging function may support link-layerencryption/decryption on the bridged data. Encryption/Decryption may besoftware only or hardware assisted depending upon the desiredperformance of the system. In either case, this function will execute asan application on top of pSOS.

Another component of the LANU software 620 is the HDT LAN manager 622.The HDT 12 LAN 591 is used to communicate system messages between theelements of the HDT 12. During pre-provisioning, the SCNU 58 willcommunicate system parameters such as CRY, IDT ID, and number ofchannels accessible by the CDM to the mLANU 580. These parameters willbe used in the construction of the call provisioning table resident onthe mLANU. During call provisioning, the mLANU will examine theprovisioning table for available DSOs and use the HDT LAN 591 to set upMARIO configurations on other LANUs 580.

Another important function of the HDT LAN manager software 622 issupport for field software upgrades. During download, the LANU 580 willtake the image from the HDT LAN and store it in on-board FLASH memory.Aside from the SCNU 58, the LANU will be the only board in the HDT 12that will load its image from its own FLASH on power up. As such,support for image download from the SCNU 58 Ethernet port will need tobe added to the SCNU software. A final function of the LAN managersoftware 622 is to provide the network management access to the SNMPenvironment of the LANU.

The final major task of the LANU software 620 is the data IDT 623. Asdescribed above, the system 10 of the present invention is designed toprovide access from POTS to a CO switch. As such, the burden of resourceallocation and assignment of DSO terminates at the witch. Since the dataarchitecture of the system 10 terminates at the head-end there is nosuch centralized resource in the architecture to provide the services ofthe switch. In order to provide the services required to terminate thedata “calls” a single LANU 580 functions as the data IDT.

The function of the data IDT is to provide a single point of referencefor the data resources of the HDT 12. During pre-provisioning of theLANU hardware a LANU 580 will be designated the “Master” LANU (mLANU)and assigned an IDT identifier. The mLANU 580 will then take on thefunction of the switch for data calls by maintaining a table that mapsCRVs to service level (# of channels). In addition, the mLANU willmaintain a map of all available DSOs an all LANUs (including the mLANUitself) installed at the HDT. A copy of the call provisioning table willbe kept in on-board FLASH so it can survive a power loss.

In order to maintain compatibility with standard telephony traffic, themeans of communication between the CTSU 54 and mLANU will be a TMCconnection over one of the DS0s within a MARIO. During callprovisioning, the CTSU 54 sends a setup message over the TMC and themLANU will respond with a “Make Cross Connect” message that identifiesthe DSO, DS1, and CRY for the connection. As discussed previously, themLANU will also configure the LANU 580 for the connection throughcommunications over the HDT 12 LAN 591. Therefore the data IDT software623 will emulate the switch though its communications with the CTSU 54over the TMC using Q.931 compatible messaging.

On all LANUs in the HDT 12, whether master or not, the data IDT software623 will be responsible for configuring the TSA and communicating withthe bridging task. In configuring the TSA, the data IDT will monitor thesequence numbers in the ninth bit signaling and appropriately configurethe interconnect so that the order within a multichannel call ismaintained. In addition, the data IDT software 623 will communicate thestate of the connection to the bridging task to open up the data pipe.

Another important function of the data IDT software 623 is to providethe information needed to provide billing and other accountingfunctions. As an IDT like function the data IDT will also supportstandard CMISE objects.

CDM Software

Software provided in a CDM 535 (executing on the local processor 605 or614 and represented by such elements) will provide the same types offunctions. The major function of the CDM software is to provide thebridge/router (brouter) functionality at the customer-end. In supportingthe brouter function the CDM software supports IP routing, PPP, andSLIP. As part of IP support the CDM supports TFTP for downloading newcode images. The CDM also supports a standard SNMP agent with a fullMIB-II information base. Preferably, the software executes on either the68302 (605) or 68360 (614) processor.

The control of the modem portion is with the standard MISU or HISUsoftware running on a Motorola 68HC11. This code supports all the alarmconditions and communications set out above for IOC communications. Theinterface between the RF modem and the brouter is preferably a hardwareonly implementation.

Network Management

Network management of the data architecture of system 500 is preferablyprovided by both CMISE and SNMP. The CMISE portion of network managementwill be responsible for the transport mechanism for data, while SNMPwill be used for data network oriented management. In this environment,SNMP is an overlay to the CMISE environment.

As with all telephony services, the data architecture will depend uponCMISE for network management of call provisioning and other transportrelated functions. In addition, CMISE will be responsible for accountingon data connections. This approach provides for a very flexible billingsystem where services can be billed per connection time, bytes passed,or packets passed. Statistics will be collected in the mLANU andreported to the network manager.

SNMP management is used to provide data services management for the dataarchitecture. In this way the data architecture will resemble a standarddata network. Within SNMP management, the LANU 580 and CDM will maintainSNMP agents compliant with the MIB-II standard. In order to support anSNMP agent both the LANU and the CDM will need to support the UDP and IPprotocols in addition to the SNMP protocol. In order to provide a singlepoint of management for data and telephony, both CMISE and SNMP arepreferably integrated into the same element manager. This level ofintegration will simplify billing by providing several options such asbill by connection time, bytes passed and packet passed.

Asymmetrical Data Delivery

For many casual residential users, data traffic can be characterized asmostly “bursty” (intermittent), downstream traffic with relatively,small upstream needs. The most cost effective means of deliveringservices such as Web browsing, file downloads, and CD-ROM preview isasymmetrical transport. The asymmetrical data transport embodiment ofthe invention includes a customer premise unit or Personal Cable DataModem that contains a 30 Mbps, QAM downstream demodulator (PCDM-30) 620a, as shown in FIG. 112. PCDM-30 also includes an OFDM upstreammodulator supporting a minimum of 64 Kbps guaranteed, non-sharedbandwidth. The connection to the customer-end equipment is 10BaseTEthernet that supports standard TCP/IP.

At the head end 32, an ASMU 622 a supports multiple users on a single 30Mbps channel which occupies 6 MHz of spectrum outside of the channels ofthe telephony transport system 10. In addition to the downstreammodulator, the ASMU 622 a concentrates the return channels byinterfacing with the HDT 12. Upstream traffic is carried over as asingle DSO and integrated with the downstream transport on the ASMU 622a. The connection from the ASMU to the head-end services is 10BaseT, buthigher capacity industry standard connections are also possible.

The ASMU 622 a sits at the head end 32, but not in the HDT 12. Thefunction of the ASMU 622 a is to integrate the upstream path for up to400 DSOs (Configurable from 64 Kbps to 512 Kbps) and a 30 Mbps shareddownstream. Each LANU 580 will generate an 8.2 Mbps HDLC stream thatcontains the 64 Kbps Ethernet packets from all the users that areattached to LANU through the transport system. On the ASMU 622 a, up tofour of these, are aggregated, and sent out to the head services over10BaseT Ethernet. In the downstream direction, the data on the 100BaseTis filtered on the ASMU, and those packets destined for the customer endproducts are accepted and then modulated onto the 30 Mbps shared medium.

In order to register a modem, the customer-end modem sends out an IPpacket to identify itself. This causes the LANU 580 to assign an HDLCaddress that is mapped to the MAC address of device. This information ispassed to the ASMU 622 a so that the HDLC address can be used by themodulator over the HDT 12 backplane LAN 591. The HDLC address andfrequency for the tuner is also sent to the customer-end over thedownstream telephony path and registered at the customer end. Thisaddress is then used by the customer-end equipment to filter the 30 Mbpsdownstream channel.

One advantage of the asymmetrical system is that a relatively largenumber of casual users (300+) can be supported by a single multi-megabitdownstream transport, with an optimal amount of upstream capacity. Theimplementation of the downstream matches the downstream of other cabledata modems in use and additionally provides superior, high capacityupstream. Since casual users place lesser demands on the network (peakutilization is lower than that of business), users can be concentratedon the return channel, thus lowering head-end 32 costs.

The upstream channel in asymmetrical applications is still important dueto the nature of the acknowledge protocol of TCP/IP, where blocks ofdata sent in the downstream must wait on an acknowledge message from thereceiver before subsequent data blocks are sent. If the upstream channelis too small, the downstream channel will stall, reducing theutilization of the downstream bandwidth. By guaranteeing a minimum of 64Kbps to each user, the asymmetrical system can deliver greater than 1Mbps sustained to each user, matching the capacity of most residentialcomputer equipment. Another advantage is the superior security of OFDMin the upstream. Unlike other shared upstream modem products currentlyavailable, the asymmetrical system herein described preventsinformation, such as bank accounts and credit card numbers exchangedduring on-line Internet shopping, from being “seen” by other modems onthe network.

Summary of Data Delivery System Advantages

Thus, the symmetrical embodiment of system 500 provides many options forthe delivery of data services over HFC distribution network 11 to theresidence or business. The DMSM 550 provides from 64 Kbps to 512 Kbpsaccess to head-end resources over a 10BaseT connection or RS232 (64 Kbpsservice). The service is symmetrical (same data rate upstream anddownstream), non-shared and dedicated to each user, providing aguaranteed level of service. As an add-in card to the HISU, the DMSM 550provides complete transport integration with telephony, supplyinghigh-speed data and two POTS lines to the residence. The PCDM 540provides the same data transport capabilities as the DMSM 550 in astandalone configuration, packaged in a traditional modem housing. Thisimplementation is ideal for premises or installations where telephony isnot deployed.

The DMCU 560 is an MISU channel unit that provides higher data ratesthan either the DMSM or PCDM-512K. The DMCU 560 router manages foursubscribers who share up to 8.192 Mbps of symmetrical bandwidth. Therouter implementation guarantees that all four subscribers on the DMCU560 have private connections. The DMCU 560 works well for multipledwelling installations for Internet access and small businessconnections where symmetrical, non-shared data access is required.

At the head end, the LANU 580 provides the concentration of up to 100DSOs in flexible combinations of various data rates, from 641 Kbps to512 Kbps for residential, and up to 8.192 Mbps for business applicationson a single, industry standard 10BaseT connection. An HDT 12 can beconfigured with up to seven LANUs, concentrating up to 700 DSOs. Inaddition to the industry standard transparent bridging function, theLANU also provides the intelligence for the dynamic, adaptive allocationof bandwidth capacity to optimize transport during times of heavyloading. This capability enables an HFC service provider using system 10to mix residential and business data services in a single 6 MHz channelwithout compromising the quality of service for business connectionsduring peak Internet access times. Dynamic-allocation allows thecustomer units to efficiently utilize the data transport by droppingconnections at times of no traffic and re-establishing them when data isready to send. Each time a connection is established the LANU 580 willallocate bandwidth of up to a maximum of 512 Kbps, depending upon thenetwork load, with a minimum of 64 Kbps. Finally, the LANU collectsdetailed traffic statistics that can be used for a variety of billingmethods, for instance bill by connect time.

System 500 is particularly effective in meeting the special needs andhigher expectations of business applications. Businesses tend to requirea higher level of upstream signaling in order to support applicationssuch as telecommuting and videoconferencing. Most cable data modemnetwork architectures can provide only limited upstream capacity, butADC is able to, offer a very high capacity upstream due to theefficiency of OFDM and frequency agility. Guaranteed bandwidth is ofequal importance to upstream capabilities. Businesses must have fullaccess to their pipeline at all times, regardless of other traffic onthe network. With system 500, once a premium user's bandwidth has beenestablished, it cannot be diminished, regardless of the number of userswho subsequently access the network.

The security of the data being transported is also a major concern tobusinesses. Security at the transport layer (encryption and secure keyexchange) and at the network layer (filtering) is provided by currenttransport technologies. System 500 also provides additional security atthe physical layer, made possible by utilizing frequency scramblingwithin the OFDM transport.

The symmetrical product line is well suited for “power” Internet userswho use their PC's not only for casual Web browsing but for remote LANaccess, telecommuting, real-time audio, and possibly videoteleconferencing. While these users are demanding, they are frequentlyearly adopters of technology who will push the limits of Internet accessand Internet applications, making the symmetrical, non-shared,guaranteed quality of service of the symmetrical products a requirement.

For both residence and business users, the symmetrical embodiment ofsystem 500 provides for superior integration with telephony. Byutilizing OFDM transport in both the upstream and downstream, thesymmetrical system can carry data in the same 6 MHz channel as telephonytraffic. This capability is ideal for smaller installations and earlydeployment where efficient use of spectrum is important. In addition,OFDM provides a very secure data delivery stem by implementing apoint-to-multipoint bridge for data where two customer premises unitsnever share the same digital data stream. The delivery of data oversystem 500 requires the efficient allocation of available bandwidth andnetwork management of system resources.

System 500 provides a completely scalable data architecture bydynamically allocating bandwidth for data traffic through itsutilization of a subset of standard TR303N5 call processing software.This system capability gives HFC service providers the flexibility totailor the configuration of head-end resources to satisfy the diverseneeds of their subscriber base. Subscriber services can be provisionedat the head end as symmetrical fixed, symmetrical variable, orasymmetrical services. As the subscriber mix changes or subscribersupgrade service, head-end resources can be re-provisioned to meet thenew requirements. For example, users can be easily reconfigured toupgrade from 64 Kbps service to 512 Kbps or even from asymmetrical tosymmetrical. For capacity planning, data bandwidth is allocated as anumber of DSOs to potential users with a single HDT supporting up to 720DSOs. The number of users supported is then a function of service level(number of DSOS) and concentration ratio (number of users per DSO).

To ensure that service providers have an effective tool to manage theircable data networks, system 500 offers an integrated data/telephonynetwork management solution. Data management is based on industrystandard SNMP agents and MIBs (management information bases), which arethen combined into an integrated data/telephony network managementenvironment. Integration of data delivery and telephony into a singlenetwork management system has several advantages:

-   -   (1) Symmetrical data, asymmetrical data, and telephony elements        can be managed by the same element manager.    -   (2) Less support staff is required.    -   (3) Better integration with billing.    -   (4) Better fault isolation.    -   (5) Lower Mean Time To Repair (MTTR).

Thus, system 500 provides a single, integrated system that can meet thediverse needs of potential subscribers, from casual Internet browsers tohigh-capacity business users. The integrated solution gives HFC serviceproviders a single point of network management that results in reducedsupport costs, reduced staffing costs, and shortened time to turn-up newservices. Finally, the OFDM technology of system 500 provides data,video and telephony services in a bandwidth-efficient system thatreduces the demands on a very valuable commodity for HFC serviceproviders spectrum.

ATM Embodiment

The system 500 of the present invention can also be configured to carrydata from an Asynchronous Transport Mode (ATM) network. As shown inFIGS. 114 and 115, system 10 or 500 of the present invention is modifiedto include an ATM multiplexer/modulator 650 which can receive ATM datafrom an ATM network 652 and modulate it onto the HFC network. In onepreferred embodiment, digital video data is delivered over ATM network652, multiplexed and modulated using multiplexer/modulator 650 onto theHFC network in RF digital OFDM format on assigned data and/or telephonychannels between the head end and a subscriber, as for example describedabove with respect to system 10 or 500. A digital set top box 654receives the digital video, formatted for example in 4.0 Mbps MPEG orequivalent, and converts it to video for display on a television 656. Areturn path to the HDT 12 over a telephony or data channel allows forinteractive digital video. A video server 658 and ATM switch 660,feeding the ATM multiplexer/modulator 650, is shown in FIG. 115.

Embodiment of Control Aspects of Telecommunications System ChannelManager

In one embodiment, communication system 10 of FIG. 1 includes channelmanager 900 of FIG. 59 to control various aspects of the dynamicallocation of channels to ISUs 100. For example, channel manager 900assigns each ISU 100 to a subband, allocates channels in the subband toan ISU to complete a communication link, and monitors the channel todetect and avoid use of corrupted channels. Channel manager 900implements further functions as described below to coordinate the use ofthe channels in a 6 MHz transmission channel to ISUs 100.

Channel manager 900 may comprise software executed by a processorresident in each CXMU 56 of each HDT 12. Channel manager 900 receivesevents from board support software 902, IOC and modem communicators 904,ISU ranger 906, and administrator 908. Channel manager 900 also sendsmessages to IOC and modem communicators 904 for allocation orreallocation of channels. Channel manager 900 uses two types of channelsto communicate control data to the ISUs. First, channel manager 900broadcasts control data over the IOC channels to the ISUs. The controldata on the IOC channels contains an identification signal thatindicates the ISU to receive the control data. Further, channel manager900 uses an ISU demand link channel, referred to as an IDL channel, fornon-time-critical transport of data between head end 32 and an ISU whenthe data is of a size that would benefit from a transmission channelwith more bandwidth than the IOC. Typically, the data rate for the IOCchannel is 16 Kbps and the data rate for the IDL channel is 64 Kbps dueto the amount of data contained in each package or frame. Typically,control signals contain four data bytes or less per frame or package.The IDL channel is used to transmit data packages that are larger thanthis. For example, the IDL channel is used to download software to anISU, provision a channel unit, transmit future channel unit functions,or transmit protocols. In one embodiment, HDT 12 only implements one IDLat a time. The IDL channel is described in more detail below.

Subband Assignment and Channel Allocation

Channel manager 900 is responsible for assigning an ISU to a subband andfor allocating payload channels for communications links to the ISU.Appropriate selection of subband and payload channel improve theperformance of communication system 10. Channel manager 900 furthermonitors the channels and reassigns subbands and reallocates channels asnecessary to maintain acceptable communications links between head end32 and ISUs 100.

Channel manager 900 selects a subband for an ISU in severalcircumstances: during acquisition, when an HISU is assigned to a subbandthat has insufficient payload channels to meet a request, and during anHISU IOC timeout event. An IOC timeout event occurs when acknowledgmentsare not received by channel manager 900 from an ISU within a specifiedtime period. With a timeout, it is assumed that the downstreamcommunications to the ISU are still in tact even though the upstreamcommunications have become corrupted due to noise or collisions. Thus, amessage on the IOC to retune to a new subband is assumed to reach theISU despite the lack of an acknowledgment.

In each case in which an ISU is assigned to a subband, channel manager900 uses various criteria to select the subband for an ISU. FIG. 62 is aflow chart that illustrates one embodiment of a method for assigning anISU to a subband. According to this method, channel manager 900 firstselects a subband 6202. Channel manager 900 then determines whetheraddition of the ISU to the subband would provide an acceptable load onthe IOC channel 6204. For example, channel manager considers the numberof ISUs assigned to a subband. Further, channel manager considers thetype of ISU and the likely load that the ISU will place on the IOCchannel. By considering these factors, channel manager 900 canselectively distribute the load on the IOC channels so as to facilitatetimely communication of control data to and from the ISU. This alsoallows channel manager 900 to evenly distribute the ISUs over theavailable subbands such that a like number of ISUs occupy each subband.Channel manager 900 also weighs the number of available channels 6206within the subband and their transmission quality 6208 as recorded inthe tables of channel manager 900. Channels with longer low-error ratehistories will be used first. Channels previously marked bad andreallocated for monitoring will be used last. Based on these criteria,channel manager selects a subband for each ISU 6210.

FIGS. 63, 64 and 65 are frequency spectrum diagrams that illustrateinitial assignment of HISUs and MISUs to various subbands in a 6 MHztransmission channel. These Figures show that channel manager 900attempts to evenly distribute the ISUs across the transmission channel.As depicted in FIG. 63, channel manager 900 begins assigning subbands atthe middle of the 6 MHz transmission channel. Channel manager 900 thenmoves out toward the ends of the transmission channel. For example, thefirst HISU is assigned to subband number 12 and the twenty-fourth HISUis assigned to subband 0. It is noted that more than one ISU can beassigned to a subband. As depicted in FIG. 64, channel manager 900initially assigns the first MISU to subbands 0 through 12 and the nextMISU to subbands 11 through 23. As depicted in FIG. 65, when HISUs andMISUs are assigned to the same subbands, channel manager assigns thesubbands so as to evenly distribute the ISUs over the availablesubbands. It is noted that the factors listed for use in selecting asubband are shown by way of example and not by way of limitation. Otherfactors can be added and the weight given to each factor can be adjustedwithout departing from the spirit and scope of the present invention.

FIG. 60 is a flow chart that illustrates one embodiment for a method forallocating payload channels in a subband by channel manager 900. Channelmanager 900 attempts to maintain an acceptable distribution of bandwidthwithin a subband to reduce the need for reallocation of payload channelswithin the subband. Further, the goal is to allocate channelsappropriately across the 6 MHz transmission channel to avoid having toreallocate channels that are currently in use. A channel can beallocated to an ISU only from the available channels in the subband towhich the ISU is assigned.

Channel manager 900 receives a request for allocation of a payloadchannel from either the SCNU 58 or CTSU 54. At block 912, channelmanager 900 decides whether sufficient payload channels are available inthe current subband to fulfill the request. If sufficient channels areavailable, the method proceeds to block 914 and determines whether oneof the available channels is the IDL channel. If the IDL channel is notone of the available channels, channel manager 900 allocates a channelfor each channel requested by CTSU 54 or SCNU 58 at blocks 916 and 918.Channel manager 900 selects the channels based on several criteria thatincrease the likelihood of achieving a connection with acceptablequality levels. For example, channel manager 900 can use the methodshown in FIG. 61. According to this method, channel manager 900 beginsthe selection process 6100 by identifying available payload channels6102 that are located toward the center of the 6 MHz transmissionchannel. Typically, channels that are nearer to the edge of the 6 MHzchannel exhibit higher bit error rates than the channels that are closerto the center. Further, channel manager 900 can also considerlimitations of the ISU and the requested service in selecting a payloadchannel. For example, the ISU may be preset for use only with odd oreven payload channels. This information may be included in a ROM on theISU and provided to the channel manager when channel allocation isrequested or during acquisition. Further, channel manager 900 uses dataon the quality of transmissions over the identified channels stored intables in channel manager 900 to determine which available payloadchannels have an acceptable error history 6104, for example, bit errorrate. Other appropriate criteria can be used in channel selection thatalso tend to increase the chances of producing a connection withacceptable quality 6104. Based on these criteria, channel managerselects a payload channel to allocate to the ISU 6106.

If, at block 914, channel manager 900 determines that one of theavailable channels is the IDL channel, channel manager 900 deallocatesthe payload channel allocated to be the IDL channel at blocks 920 and922 due to the lower priority of communications over the IDL channel.

If, at block 912, channel manager 900 determines that sufficient payloadchannels are not available in the current subband, channel manager 900determines whether the request is for an HISU 68 or an MISU 66 at block924. If the request is for an MISU 66, channel manager 900 sends amessage to the requestor that the request has failed at block 926.

If, at block 924, channel manager determines that the request is for anHISU, then channel manager 900 selects a different subband at block 928by weighing the criteria as described above with respect to selecting asubband. Channel manager 900 further marks the channels unavailable inthe new subband at block 930 and deallocates channels allocated to theISU in the prior subband at block 932. At block 934, channel manager 900assigns the new subband and proceeds to allocate channels as necessaryat blocks 916 and 918.

An example of reassigning an ISU to a new subband to accommodate arequest for a payload channel is shown in FIGS. 66 and 67. In thisexample, ISUs A, B, C, and D are initially assigned to subband 4 andISUs E, F, and G are assigned to subband 17 as depicted in FIG. 66. Insubband 4, all payload channels except payload channel 0 are allocated.In this case, channel manager 900 receives a request for two payloadchannels for ISU C. Since only one payload channel is available, channelmanager 900 reassigns ISU C to subband 17 which has sufficient payloadchannels available to handle the current load of ISU C plus theadditional two payload channels as shown in FIG. 67.

Channel Reallocation

Channel monitoring and allocation or reallocation based thereon may beused to avoid ingress. External variables can adversely affect thequality of a given channel. These variables are numerous, and can rangefrom electromagnetic interference to a physical break in an opticalfiber. A physical break in an optical fiber severs the communicationlink and cannot be avoided by switching channels, however, a channelwhich is electrically interfered with can be avoided until theinterference is gone. After the interference is gone the channel couldbe used again.

Channel Monitoring

Channel monitor 900 monitors the payload channels for errors to help indetermining which channels are acceptable for transmission for specificservices. One input to channel manager 900 is parity errors which areavailable from hardware per the DSO+ channels; the DSO+ channels being10-bit channels with one of the bits having a parity or data integritybit inserted in the channel as previously discussed. The parity errorinformation on a particular channel is used as raw data which is sampledand integrated over time to arrive at a quality status for that channel.In one embodiment, parity errors that are detected on downstream payloadchannels are communicated to head end 32 over an associated upstreamchannel. When the error is detected in the downstream transmission, theparity bit for the upstream transmission is corrupted by intentionallysetting the parity bit to the wrong value to indicate the incorrectparity in the downstream transmission path. Thus, the ISU informs thehead end of errors in the downstream path.

To monitor the payload channels, channel manager 900 needs an activeupstream signal on each payload channel. However, at any given time,some payload channels may not be allocated and some allocated channelsmay not be active. Thus, these payload channels do not provide thenecessary upstream signals to the head end to monitor the quality of thepayload channels. To compensate for these idle and unallocated payloadchannels, channel manager 900 places these channels in loop back mode tomonitor the quality. In this case, channel manager 900 sets up thepayload channel, transmits data to the ISU on the payload channel andthe ISU transmits back specified data on an associated upstream payloadchannel. Channel manager 900 monitors these channels at the head end todetermine error rates for the channels. Thus, the unallocated or idlepayload channel can be monitored for errors the same as with activechannels. The goal of channel manager 900 is to have payload establishedon all of the payload channels at a given time. However, it may beacceptable to monitor the performance of each channel at least once anhour if not active.

Channel manager 900 randomly selects and uses ISUs to monitor payloadchannels in loopback mode described above. This provides severalbenefits to the system. First, this allows channel manager to handle thediverse layout of a cable plant. Channel manager 900 sets up and usespaths over different legs from the various ODNs of the system. Further,random cycling of the ISUs used in the loop back mode allows the systemto properly distribute power in the coaxial network. Specifically, thisrandom selection of ISUs for loopback mode applies to concentration typeservices.

As described below, some ISUs are powered down when not active. When theISU is powered down, the upstream modem at the head end detects thiscondition and sends a specified signal to the CXMC so that channelmanager does not use the ISU for loop back purposes. Thus, powered downISUs do not produce unnecessary errors.

FIG. 68 is a flow chart that illustrates a method for monitoring payloadchannels by channel manager 900. Channel manager 900 reads parity errorregisters 901 of the CXMU 56 are read every 10 ms. Generally, the errorcounts are used to update the channel quality database and determinewhich (if any) channels require reallocation. The database of channelmanager 900 contains an ongoing record of each channel 903. Anaccumulator sums the errors 905 with previously recorded errors toupdate the database. The database organizes the history of the channelsin categories such as: current ISU assigned to the channel, start ofmonitoring, end of monitoring, total error, errors in last day, in lastweek, number of seconds since last error, severe errors in last day, inlast week, and current service type, such as ISDN, assigned to thechannel. When the channel is a regular (non-loop back) payload channel907, channel manager 900 determines whether the performance statisticsin the database are within service specific threshold 909. When thestatistics unacceptably exceed the threshold 910, channel manager 900reallocates the channel 911 using a “make before break” procedure toreduce the disruption from reallocating the channel. Thus, channelmanager 900 allocates the new payload channel for the connection beforedeallocating the current payload channel.

Two issues presented by periodic parity monitoring as described abovemust be addressed in order to estimate the bit error rate correspondingto the observed count of parity errors in a monitoring period todetermine if a channel is corrupted. The first is the nature of parityitself. Accepted practice for data formats using block error detectionassumes that an errored block represents one bit of error, even thoughthe error actually represents a large number of data bits. Due to thenature of the data transport system, errors injected into modulated dataare expected to randomize the data. This means that the average erroredframe will consist of four (4) errored data bits (excluding the ninthbit). Since parity detects only odd bit errors, half of all erroredframes are not detected by parity. Therefore, each parity (frame) errorinduced by transport interference represents an average of 8 (data) bitsof error. Second, each monitoring parity error represents 80 frames ofdata (10 ms/125 us). Since the parity error is latched, all errors willbe detected, but multiple errors will be detected as one error.

The bit error rate (BER) used as a basis for determining when toreallocate a channel has been chosen as 10⁻³. Therefore, the acceptablenumber of parity errors in a one second interval that do not exceed 10-3must be determined. To establish the acceptable parity errors, theprobable number of frame errors represented by each observed (monitored)parity error must be predicted. Given the number of monitored parityerrors, the probable number of frame errors per monitored parity error,and the number of bit errors represented by a frame (parity) error, aprobable bit error rate can be derived.

A statistical technique is used and the following assumptions are made:

-   -   1. Errors have a Poisson distribution, and    -   2. If the number of monitored parity errors is small <<10) with        respect to the total number of “samples” (100), the monitored        parity error rate (MPER) reflects the mean frame error rate        (FER).

Since a monitored parity error (MPE) represents 80 frames, assumption 2implies that the number of frame errors (FEs) “behind” each parity erroris equal to 80 MPER. That is, for 100 parity samples at 10 ms persample, the mean number of frame errors per parity error is equal to 0.8times the count of MPEs in one second. For example, if 3 MPEs areobserved in a one second period, the mean number of FEs for each MPE is2.4. Multiplying the desired bit error rate times the sample size anddividing by the bit errors per frame error yields the equivalent numberof frame errors in the sample. The number of FEs is also equal to theproduct of the number of MPEs and the number of FEs per MPE. Given thedesired BER, a solution set is illustrated below in Equation 3.

$\begin{matrix}{\left( {{MPE}\;\frac{FE}{MPE}} \right) = {0.8\mspace{14mu}{MPE}}} & \left( {{Equation}\mspace{14mu} 3} \right)\end{matrix}$

The Poisson distribution, as illustrated below in Equation 4, is used tocompute the probability of a given number of FEs represented by a MPE(X), and assumption 2, above, is used to arrive at the mean number ofFEs per MPE (p).

$\begin{matrix}{{P(x)} = \frac{e^{- \mu}\mu^{x}}{x!}} & \left( {{Equation}\mspace{14mu} 4} \right)\end{matrix}$

Since the desired bit error rate is a maximum, the Poisson equation isapplied successively with values for x of 0 up to the maximum number.The sum of these probabilities is the probability that no more than xframe errors occurred for each monitored parity error. The results for abit error rate of 10⁻³ and bit errors per frame error of 1 and 8 areshown in Table 11.

TABLE 11 Bit Error Rate Probability Maximum Average Bit Errors MonitoredFrame Errors/ Frame Errors/ Probability per Frame Parity MonitoredMonitored of Error Errors Parity Error (χ) Parity Error (μ) BER <−10³ 82 4 1.6 98% 3 3 2.4 78% 4 2 3.2 38% 1 8 8 6.4 80% 9 7 7.2 56% 10 7 8.045%

Using this technique, a value of 4 monitored parity errors detectedduring a one second integration was determined as the threshold toreallocate service of an ISU to a new channel. This result is arrived atby assuming a worst case of 8 bit errors per frame error, but aprobability of only 38% that the bit error rate is better than 10⁻³. Theproduct of the bit errors per frame, monitored parity errors and maximumframe errors per monitored parity error must be 64, for a bit error rateof 10⁻³ (64 errors in 64 k bits). Therefore, when the sampling of theparity errors in the error timer event is four or greater, the channelallocator is notified of a corrupted channel.

DSO Reordering

Some telecommunications services use multiple DSOs (payload channels) toform a communication link in communication system 10. For example, ISDNuses three DSOs to form three payload channels identified namely as B1,B2, and D. To operate properly, the DSOs typically are assigned in aspecific sequence. Once the payload channels for the service areassigned, the channel unit associated with the service expects toreceive the payload channels in a specific order. When one of thepayload channels becomes corrupted, channel manager 900 allocates adifferent DSO channel for the corrupted channel and the sequence of theDSOs is altered.

This problem could be avoided by reallocating all three DSOs. However,this is a time consuming process and could cause transient disruption ofthe service. As an alternative, channel manager 900 can assign anaddress to the DS0s when the multiple DSO service is initiated. Thisaddress can be used by the channel unit to reconstruct the order of theDSOs on the fly if one or more of the DSOs is reallocated out ofsequence with the other DSOs. For example, in the channel enable signalfrom CXMU 56 on the IOC channel, a BIC state signal can be used toidentify the correct order for each DSO. Thus, channel manager 900 canallocate the DSOs in any order and the channel unit can remap the DSOsto the correct order at the ISU. It is noted that the DSOs must still beallocated in different time slots.

ISU Data Link (IDL) Channel

The IDL is a standard payload channel that is dynamically assigned totransmit control data between HDT 12 and ISU 100 when the amount of dataexceeds the parameters of the lower bandwidth IOC channel. The IDLchannel can provide full duplex communication or simplex broadcast fromHDT 12 to one or more ISUs 100. Channel manager 900 dynamicallyallocates the IDL channel as needed for non-time critical transport ofdata as described above.

The IDL messages in both directions are variable in length. The IDL datais transmitted over HFC distribution network 11 at a rate of 64 Kbpswhich is one byte per 8 kHz frame. The IDL channel uses one of the 240payload channels and the protocol for transmitting IDL messages ishandled by the processor on CXMU 56. The processor uses cyclicalredundancy codes (CRC) and positive acknowledgments to manage errorchecking of IDL messages.

The IDL channel can be used to transmit various kinds of data. Forexample, the IDL channel can be used to provide data to an ISU toconfigure a payload channel for use with a specific protocol. Forexample, the IDL channel can be used to down load data to configure apayload channel for use with the LAPB protocol or any other appropriateprotocol, including proprietary protocols. Similarly, the IDL channelcan be used to download software to an ISU. Transmission over the IDLchannel has a lower priority than regular payload transmissions. Thus,channel manager 900 deallocates an IDL channel before completion of thedata transmission when channel manager 900 receives a request thatrequires use of the payload channel that is currently allocated to theIDL.

FIG. 69 is a flow chart that illustrates an embodiment of a method forallocating a payload channel to the ISU data link. At block 330 a,channel manager 900 receives a request for an IDL channel. At block 332a, channel manager 900 determines whether a payload channel isavailable. If a payload channel is available, channel manager 900allocates the payload channel to the IDL channel 335 and the data istransmitted to the identified ISU. If, however, a channel is notavailable, channel manager determines whether one of the allocatedchannels is idle by checking the hook state of a line of a channel unit342. If the line is on hook 339, then channel manager 900 reallocatesthe channel to the IDL channel 343 until the IDL transmission iscomplete. If however, channel manager receives a request for acommunication link to the line of the channel unit, channel managerinterrupts the IDL channel and reallocates the payload channel to thechannel unit.

Power Down

Channel manager 900 can power down an ISU during periods of non-use toreduce energy costs of communication system 10. To power down the ISU,channel manager 900 must determine that all conditions for powering downthe ISU are met. For example, channel manager 900 can determine if thelines of the channel units of the ISU provide service that can bepowered down. Such services may include, for example, analog servicessuch as POTS and COIN. Further, the lines must be idle. For example,channel manager 900 can determine if a line is idle based on the line'shook status or other appropriate criteria. Channel manager 900 checksthe status of the lines each time a line goes from off-hook to on-hook.Channel manager 900 further checks the status of the lines every secondto monitor the hook status. It is noted, however, that channel manager900 will not power down an ISU that is the monitoring ISU for a subband.

When channel manager 900 determines that an ISU can be powered down, theISUs' transmitter is disabled. Head end 32 detects the loss of power tothe ISU and sends an idle pattern upstream to the switch. An IOC controlmessage to or from the IOC will power-up the ISU. The IOC traffic to orfrom the ISU indicates to the background task in charge of powering downISUs to check the ISU against the criteria for powering down again.

It is understood that the above description is intended to beillustrative, and not restrictive. Many other embodiments will beapparent to those of skill in the art upon reviewing the abovedescription. The scope of the invention should, therefore, be determinedwith reference to the appended claims, along with the full scope ofequivalents to which such claims are entitled.

Further Embodiment of Telephony Transport System

With reference to FIG. 116, a general description of a hybrid fiber/coaxcommunications network 1006 in accordance with the present inventionshall be described. Telephony and video information from existingtelephone and video services generally shown by trunk line 1008 isreceived by and processed by head end 1010. Head end 1010 includes aplurality of host distribution terminals (HDT) 1012 for telephony datainterface and video host distribution terminal (VHDT) 1014 for videodata interface. Host distribution terminals 1012 and VHDT 1014 includetransmitters and receivers for communicating the video and telephonyinformation between the video and telephony signal distribution network1006 in accordance with the present invention and the existing telephonyand video services as represented generally by trunk line 1008.

The video information is optically transmitted downstream via opticalfiber line 1017 to splitter 1018 which splits the optical video signalsfor transmission on a plurality of optical fibers 1022 to a plurality ofoptical distribution nodes 1026. The HDT 1012 transmits opticaltelephony signals via optical fiber link 1020 to the opticaldistribution nodes 1026. The optical distribution nodes 1026 convert theoptical video signals and telephony signals for transmission aselectrical outputs via a coaxial distribution system 1007 to a pluralityof remote units 1042. The electrical downstream video and telephonysignals are distributed via a plurality of coaxial lines 1029 andcoaxial taps 1034 of the coaxial distribution system 1007.

The remote units 1042 include means for transmitting upstream electricaldata signals including telephony information from telephones 1076 anddata terminals 1073 and in addition may include means for transmittingset top box information from set top boxes 1078. The upstream electricaldata signals are provided by a plurality of remote units 1042 to anoptical distribution node 1026 connected thereto. The opticaldistribution node 1026 converts the upstream electrical data signals toan upstream optical data signal for transmission via optical fiber link1020 to the head end 1010.

The present invention shall now be described in further detail withreference to FIGS. 116-123. The first part of the description shallprimarily deal with downstream transmission and the second part of thedescription shall primarily be with regard to upstream transmission. Thevideo and telephony distribution network 1006 in accordance with thepresent invention, includes head end 1010 which receives video andtelephony information from video and telephony service providers viatrunk line 1008. Head end 1010 includes a plurality of host distributionterminals 1012 and a video host distribution terminal 1014. The HDT 1012includes transmitters and receivers for communicating telephonyinformation, such as T1, ISDN, or other data services information, toand from telephony service providers via trunk line 1008 and the VHDT1014 includes transmitters and receivers for communicating videoinformation, such as cable TV video information and interactive data ofsubscribers to and from video service providers via trunk line 1008.

The VHDT 1014 transmits downstream optical signals to a splitter 1018via video feeder optical fiber line 1017. The passive optical splitter1018 effectively makes four copies of the downstream high bandwidthoptical video signals. The duplicated downstream optical video signalsare distributed to the correspondingly connected optical distributionnodes 1026. One skilled in the art will readily recognize that althoughfour copies of the downstream video signals are created, that any numberof copies may be made by an appropriate splitter and that the presentinvention is not limited to any specific number.

The splitter 1018 is a passive means for splitting broad band opticalsignals without the need to employ expensive broad band optical toelectrical conversion hardware. Optical signal splitters are commonlyknown to one skilled in the art and available from numerous fiber opticcomponent manufacturers such as Gould, Inc. In the alternative, activesplitters may also be utilized. In addition, a cascaded chain of passiveor active splitters would further multiply the number of duplicatedoptical signals for application to an additional number of opticaldistribution nodes and therefore increase further the remote unitsserviceable by a single head end. Such alternatives are contemplated inaccordance with the present invention as described by the accompanyingclaims.

The VHDT 1014 can be located in a central office, cable TV head end, ora remote site and broadcast up to about 112 NTSC channels. The VHDT 1014includes a transmission system like that of the LITEAMP system availablefrom American Lightwave Systems, Inc., currently a subsidiary of theassignee hereof. Video signals are transmitted optically by amplitudemodulation of a 1300 nm laser source at the same frequency at which thesignals are received (that is. the optical transmission is a terahertzoptical carrier which is modulated with the RF video signals). Thedownstream video transmission bandwidth is about 54-725 MHz. Oneadvantage in using the same frequency for optical transmission of thevideo signal as the frequency of the video signals when received is toprovide high bandwidth transmission with reduced conversion expense.This same-frequency transmission approach means that the modulationdownstream requires optical to electrical conversion or proportionalconversion with a photo diode and perhaps amplification, but nofrequency conversion. In addition, there is no sample data bandwidthreduction and little loss of resolution.

Alternative embodiments of the VHDT may employ other modulation andmixing schemes or techniques to shift the video signals in frequency,and other encoding methods to transmit the information in a codedformat. Such techniques and schemes for transmitting analog video data,in addition to those transmitting digital video data, are known to oneskilled in the art and are contemplated in accordance with the spiritand scope of the present invention as described in the accompanyingclaims.

Telephony information is transmitted downstream by HDT 1012 via opticalfiber link 1020 to a corresponding optical distribution node 1026. Amore detailed block diagram of one of the HDTs 1012 is shown in FIG.117. Each HDT 1012 includes an RF modem bank 1050 which receivestelephony information via trunk line 1008. The RF modem bank 1050includes four RF modem modules 1052 and a protection modem module 1054.Each RF modem module receives telephony information, for example timedivision multiplexed channel signals from a public switched telephoneservice, via trunk line 1008 and the telephony information modulates ananalog carrier for transmission of the downstream optical telephony databy downstream optical telephony transmitter 1080 of downstream telephonyelectrical to optical converter 1064 to a corresponding distributionnode 1026. Each RF modem module includes a transceiver 1053 and providesa downstream electrical telephony signal in one of four frequencybandwidths, each bandwidth being about 6 MHz in width like that of aCATV channel. Each 6 MHz bandwidth channel transmits data at 22Mbits/sec and can provide for transmission of 8T1 digital telephonesignals; T1 being a conventional telephone signal where 24 voicechannels are sampled at an 8 kHz rate, with 8 bits per sample (each 8bit conversation sample is termed a DSO). Each of these signals from thefour RF modem modules 1052 are transmitted via coax patch cables to acombiner 1082 of downstream telephony electrical to optical converter1064 for transmission by optical transmitter 1080. Therefore, thespectrum for the downstream optical telephony data is four separated 6MHz frequency bands containing 22 Mbits/sec of data within each 6 MHzbandwidth. The four 6 MHz frequency bands, separated by a guard band asis known to one skilled in the art, are transmitted in about the 725-800MHz bandwidth.

Any number of modulation techniques may be used for transmission of thetelephony information downstream. The transmission downstream is pointto multipoint transmission using broadcast type transmission schemes.The modulation techniques utilized and performed by RF modem module 1052may include quadrature phase shift keying (QPSK), quadrature amplitudemodulation (QAM), or other modulation techniques for providing thedesired data rate. Modulation techniques, such as QPSK and QAM, areknown to those skilled in the art and the present invention contemplatesthe use of any such modulation techniques for downstream broadcasttransmission.

The electrical to optical converter 1064 includes two transmitters 1080for downstream telephony transmission to protect the telephony datatransmitted. These transmitters are conventional and relativelyinexpensive narrow band laser transmitters. One transmitter is instandby if the other is functioning properly. Upon detection of a faultin the operating transmitter, controller 1060 switches transmission tothe standby transmitter. In contrast, the transmitter of the VHDT 1014is relatively expensive as compared to the transmitters of HDT 1012 asit is a broad band analog DFB laser transmitter. Therefore, protectionof the video information, non-essential services unlike telephony data,is left unprotected. By splitting the telephony data transmission fromthe video data transmission, protection for the telephony data alone canbe achieved. If the video data information and the telephony data weretransmitted over one optical fiber line by an expensive broad bandanalog laser, economies may dictate that protection for telephonyservices may not be possible. Therefore, separation of such transmissionis of importance.

As an alternative embodiment for providing transmission of optical videoand telephony signals to the optical distribution nodes 1027 from headend 1010 as shown in FIG. 120, the HDT 1012 and VHDT 1014 can utilizethe same optical transmitter and the same optical fiber line 1016. Thesignal then is split by splitter 1018 and four split signals areprovided to the optical distribution nodes 1027 for distribution to theremote units 1042 by the coaxial distribution system 1007 as furtherdiscussed below. However, as described above, the optical transmitterutilized would be relatively expensive due to its broad bandcapabilities, lessening the probabilities of being able to affordprotection for essential telephony services.

As one skilled in the art will recognize, optical link 1020, as shown inFIG. 117, may include four fibers, two for transmission downstream fromelectrical to optical converter 1064 and two for transmission upstreamto optical to electrical converter 1066. With the use of directionalcouplers, the number of such fibers may be cut in half In addition, thenumber of protection transmitters and fibers utilized may vary as knownto one skilled in the art and any listed number is not limiting to thepresent invention as described in the accompanying claims.

RF modem bank 1050 includes a protection RF modem module 1054 with atransceiver 1053 connected to combiner 1082 of electrical to opticalconverter 1064. Protection RF modem module 1054 is further coupled tocontroller 1060. When a fault is detected with regard to thetransmission of one of the RF modem modules 1052, a signal is generatedand applied to an input 1062 of controller 1060. Controller 1060 isalerted to the fault and provides appropriate signaling to switch theprotection RF modem module 1054 for the faulted RF modem such that theprotection RF modem module 1054 transmits within the 6 MHz bandwidth ofthe faulted RF modem module 1052 so that the four 6 MHz bandwidth signaltransmission is continued on optical fiber link 1020. The use of oneprotection RF modem module 1054 for four RF modem modules 1052 is onlyone embodiment of the present invention and the number of protection RFmodem modules relative to RF modem modules may vary as known to oneskilled in the art and described in the accompanying claims. As shown inFIG. 123, RF modem bank 1050 may include one protection module 1054 foreach RF modem module 1052. In this embodiment, the RF modem bank 1050includes three RF modem modules 1052 and three protection modules 1054for one-to-one protection.

An optical distribution node 1026 as shown in FIG. 118 receives both thedownstream optical telephony signal and the split downstream opticalvideo signal. The downstream optical video signal is applied by theoptical fiber 1022 from splitter 1018 to a downstream video receiver1120 of optical distribution node 1026. The optical distribution node1026 further includes downstream telephony receiver 1121 for receivingthe downstream optical telephony signal on optical link 1020. Theoptical video receiver 1120 utilized is like that available in theLITEAMP product line. The converted signal from video receiver 1120,proportionally converted utilizing photodiodes, is applied to bridgeramplifier 1127 along with the converted telephony signal from downstreamtelephony receiver 1121. The bridging amplifier 1127 simultaneouslyapplies four downstream electrical telephony and video signals to diplexfilters 1134. The diplex filters 1134 allow for full duplex operation byseparating the transmit and receive functions when signals of twodifferent frequency bandwidths are utilized for upstream and downstreamtransmission. There is no frequency conversion performed at the opticaldistribution nodes with respect to the video or downstream telephonysignals as the signals are passed through the optical distribution nodesto the remote units via the coaxial distribution system in the samefrequency bandwidth as they are received.

After the optical distribution node 1026 has received downstream opticalvideo signals via optical link 1022 and downstream optical telephonysignals via optical link 1020 and such signals are converted todownstream electrical video and telephony signals, the four outputs ofthe optical distribution nodes 1026 are applied to four coaxial cables1029 of coaxial cable distribution system 1007 for transmission of thedownstream electrical video and telephony signals to the remote units1042; such transmission occurs in about the 725-800 MHz bandwidth fortelephony signals and about the 54-725 MHz bandwidth for the downstreamelectrical video signals. Each optical distribution node 1026 providesfor the transmission over a plurality of coaxial cables 1029 and anynumber of outputs is contemplated in accordance with the presentinvention as described in the accompanying claims.

As shown in FIG. 116, each coaxial cable 1029 can provide a significantnumber of remote units with downstream electrical video and telephonysignals through a plurality of coaxial taps 1034. Coaxial taps arecommonly known to one skilled in the art and act as passivebidirectional pick-offs of electrical signals. Each coaxial cable 1029may have a number of coaxial taps connected in series. In addition, thecoaxial cable distribution system may use any number of amplifiers toextend the distance data can be sent over the coaxial portions of thenetwork 1006.

The downstream electrical video and telephony signals are provided fromthe coaxial taps to the remote units 1042 in a number of different ways.In one embodiment, the signal from the coaxial tap 1034 is provided to ahome integrated service unit 1070 as shown in FIG. 119. The homeintegrated service unit 1070 of FIG. 119 includes a power tap 1099coupled to a conventional power supply and ring generator 1101. Thedownstream electrical video and telephony signals are provided to a tap1097 for application of the signals to both diplex filter 1110 andingress filter 1098. The downstream video signal is provided fromingress filter 1098 to video equipment 1072 via set top box 1078. Thedownstream telephony signal is applied from diplex filter 1110 to RFdemodulator 1104 of RF modem module 1102 and the demodulated signal isapplied to an applicable service interface for processing and connectionto user equipment. For example, the RF demodulated signal is processedvia Plain Old Telephone Service (POTS) service interface 1112 for outputon twisted pairs 1118 to telephone 1076 by POTS connection 1114. Theother service interfaces such as ISDN interface or a T1 interfaceperform their conventional functions as are known to those skilled inthe art for transmittal of such information on outputs thereof to userequipment.

Ingress filter 1098 provides the remote unit 1042 with protectionagainst interference of signals applied to the video equipment 1072 asopposed to those provided to other user equipment such as telephones orcomputer terminals. Filter 1098 passes the video signals; however, itblocks those frequencies not utilized by the video equipment. Byblocking those frequencies not used by the video equipment, straysignals are eliminated that may interfere with the other servicesprovided by the network to at least the same remote unit. The set topbox 1078 is an optional element in the network 1006. It may include anadditional modem for sending interactive data therefrom back to head end1010 at frequencies unused by the video and telephony transmissions.Upstream transmission of such data is further discussed below.

Depending on the modulation processing techniques utilized at the headend 1010, the RF demodulator 1104 would include circuitry capable ofdemodulating the modulated signal. For example, if QPSK modulation isutilized then the demodulator would include processing circuitry capableof demodulating a QPSK modulated waveform as is known to one skilled inthe art.

In another embodiment of providing downstream electrical video andtelephony signals from the coaxial taps 1034 to remote units 1042, asshown in FIG. 116, a separate coaxial line from coaxial tap 1034 isutilized to provide transmission of the signals therefrom to set top box1078, and thus for providing the downstream video signals to videoequipment unit 1072. In such a case, a second coaxial line from coaxialtap 1034 would be utilized to provide the downstream telephony signalsto a multiple integrated service unit (MISU) 1044 which would be muchlike the home integrated service unit 1070 as described with regard toFIG. 119 except lacking an ingress filter 1098 and tap 1097. Unlike homeintegrated service unit 1070, the MISU 1044 would be utilized to serviceseveral remote units 1042 with telephony services via various serviceinterfaces. Whether the video and telephony signals are provided to thecurb with use of the MISU 1044 or whether the video and telephonysignals are provided directly to a home integrated service unit isstrictly one of application and either can be utilized with regard tothe same or different coaxial taps 1034 and within the same or differentcoaxial distribution systems 1007.

In addition, an optional network interface device (NID) 1075 is utilizedin the connection of telephone services to the remote units 1042,whether they are homes or businesses, as is known to those skilled inthe art and as shown in FIG. 116. The NID is generally shown by block1070 representing the home integrated service unit but is not shown inthe detail of FIG. 119. The NID performs service functions for thetelephone service provider such as looping back signals to the serviceprovider that reach the NID so as to indicate whether a failure hasoccurred somewhere in transmission to the NID or in connections from theNID to the user equipment when a failure is reported to the serviceprovider.

The above description primarily involves the downstream transmission ofvideo and telephony information from head end 1010 to remote units 1042.The upstream transmission of interactive data from set top boxes 1078and other data, for example telephony from telephones 1076, shall now bedescribed with reference to FIGS. 116-123. The description shall belimited to transmission from remote units via home integrated serviceunits as transmission from an MISU is substantially similar and easilyascertainable from such description. Home integrated service unit 1074provides set top box information from set top box 1078 and telephonyinformation from the service interfaces 1112, including information fromtelephone 1076, to the optical distribution node 1026 connected theretoby the same coaxial path as for the downstream communication. The settop box signals are transmitted by a separate RF modem of the videoservice provider at a relatively low frequency in the bandwidth of about5 to 40 MHz which is unused by telephony and video services. Thetelephony signals are also transmitted upstream in the 5-40 MHzbandwidth, usually from 10 MHz to 30 MHz. This 5-40 MHz bandwidth isreused in each coaxial path 1029 from each remote unit 1042 to therespectively connected optical distribution node 1026. As such, upstreamelectrical telephony data signals from the remote units are transmittedat the same reused frequency bandwidth of 5-40 MHz on each coaxial line1029 for input to the optical distribution node 1026. Therefore, asshown in FIG. 118, four upstream electrical telephony signals, each inthe 5-40 MHz bandwidth, are input to optical distribution node 1026, viathe respectively connected coaxial cables 1029.

The upstream transmission from an integrated service unit for multipointto point transmission utilizes time multiplexing techniques, althoughany of a number of multiple access techniques known to those skilled inthe art are contemplated in accordance with the present invention. Allthe remote units are designated a time slot for transmission. In such acase each remote unit must transmit at a particular time to maintainmultiple access with the timing being supplied using data on thedownstream paths. The upstream data is transmitted on a bit-by-bitbasis. With each remote unit assigned a time slot, the RF modem 1102 ofthe unit knows that it will not interfere with the others because it hasdetermined the time delay for each one of them and each RF modem 1102 issignaled to transmit at a precise time. Due to the high volumes ofmultiplexed serial data from several outlining remote stations andlimited bandwidth for transmission, short pulse durations are requiredfor better resolution of the data transmitted to the head end 1010.Although the data modulates a carrier and is transmitted in the 5 to 40MHz bandwidth by RF modulator 1108, because of the limited bandwidth inthe upstream direction, a pulse shaping network at each remote unit isused to generate raised cosine pulses for the rectangular or square wavebit-by-bit stream of data transmitted along the coaxial cable in thecoaxial network.

An optimal pulse shape for transmission in a band limited coaxial cablenetwork is determined by the use of Fourier calculations with a givenset of boundary conditions. Also, the Fourier calculations implement aspectral limitation constraint for the purposes of limiting the spectralcontent of the optimal pulse shape. Limiting the spectral content of thepulse shape serves two functions. The first function is to limit thespectral characteristics of the optimal pulse shape in order to preventphase dispersion at the receiving end of the transmission system. Thesecond benefit from the spectral limitation constraint is to allow theuse of relatively simple finite impulse response filters with a minimalnumber of taps.

In one embodiment of the pulse shaping network as shown in FIG. 121, 50nanosecond pulses from the RF modulator 1108 of RF modem 1102 aretransmitted to a pulse sequencer 1301 for uniform digitization. Theoutput from the pulse sequencer is then applied to a ten tapped finiteimpulse response filter (FIR filter) 1302 with associated electronics1303 to provide the addition and subtraction necessary for the filteringprocess. The output is sent to a line driver circuit for output to thecoaxial cable through diplex filter 1110. The optimal pulse waveform isa raised cosine waveform. Using such pulse shaping techniques, overcomesthe difficulty of sending extremely short pulse duration informationalong a band limited coaxial cable.

The upstream electrical telephony signals from a plurality of remoteunits, including signals from the RF modems 1102 and from modems in settop boxes 1078, are transmitted to the respectively connected opticaldistribution node 1026 as shown in FIG. 118 via the individual coaxialcables 1029. The upstream electrical signals are applied to a diplexfilter 1134 respectively connected to a coaxial cable 1029. One of thediplex filters 1134 passes the upstream electrical telephony signalapplied thereto through to combiner 1125 while the other diplex filterspass the upstream electrical telephony signals applied thereto tofrequency shifters 1128, 1130, and 1132. Frequency shifter 1128 shiftsthe upstream electrical telephony signal into the 50-85 MHz bandwidth,frequency shifter 1130 shifts another upstream electrical telephonysignal into the 100-135 MHz bandwidth and frequency shifter 1132 shiftsthe other upstream electrical telephony signal into the 150-185 MHzbandwidth. The shifted signals are combined by combiner 1125 andprovided to upstream telephony and set top control transmitters 1123.The conventional optical transmitters 1123 transmit the upstreamelectrical telephony signal as an upstream optical telephony signal tohead end 1010 via fiber optic link 1020. Once again, two transmittersare available for transmission, one in standby mode, like that in thedownstream transmission path.

The upstream optical telephony signals are received by upstreamtelephony and set top box receiver 1084 of optical to electricalconverter block 1066. The upstream optical telephony signals areconverted, split, and all the split electrical signals in the 5-40 MHz,50-85 MHz, 100-135 MHz, and 150-185 MHz are frequency shifted back tothe 540 MHz bandwidth by frequency shifters 1086, 1088, and 1090 withthe exception of the signal already in the 5-40 MHz bandwidth which ispassed through with the other frequency shifted signals from thefrequency shifters to RF switch 1094. A combined signal in the 5-40 MHzbandwidth from combiner 1092 is provided to the VHDT and the signal isprocessed for obtaining the interactive information transmitted from settop boxes 1078. The RF switch 1094 is controlled by controller 1060 andprovides the upstream telephony signals to the transceivers 1053 of thecorresponding RF modems 1052. The upstream telephony signals are thendemodulated by RF modem modules 1052 and the telephony data is providedto the service providers via trunk line 1008. The RF modem modules 1052include RF demodulator corresponding to the modulation techniquesutilized to transmit the information upstream so such information can berecovered.

As discussed previously, the controller 1060 switches protection RFmodem module 1054 for a transmitting RF modem module 1052 in thedownstream communication when a fault is detected in that module. Thecontroller also provides signaling for switching the RF switch 1094 suchthat the information which would have been provided to the faulted RFmodem module 1052 is applied to the transceiver of the protection RFmodem module 1054. Therefore, the protection modem module 1054 is thenfully within the transmit and receive loop of the system.

As shown in FIG. 122, an alternative embodiment of the present inventionincludes an optical to electrical converter 1066 wherein the receivedoptical upstream telephony signal is converted by receivers 1084 and theentire upstream electrical signal in the 5-200 MHz bandwidths is appliedto the transceivers 1053 of the RF modem modules 1052. The RF modemmodules 1052 are then operated under control of controller 1060 whichassigns the RF modem module a carrier frequency to tune to for therecovery of telephony information; the assigned frequency being afunction of the frequency shifting of the upstream signal. Theelectrical signal is still separated and frequency shifted by frequencyshifters 1086, 1088 and 1090 except for the signal already in the 5-40MHz bandwidth and then combined by combiner 1092 for application to VHDT1014.

In this embodiment, the switching of the protection modem module 1054into the system is accomplished through the controller 1060. When thecontroller 1060 detects and indicates a faulted modem module 1052, thecontroller 1060 assigns the frequency previously assigned to the faultedRF modem module to the protection module, thus establishing theprotection RF modem module 1054 fully within the transmit and receiveloop.

In another embodiment shown in FIG. 123 including one-to-one protectionfor the RF modem module, neither the RF switch used for protectionswitching for the configuration of FIG. 123 nor the additional controlrequired for protection switching for the configuration of FIG. 122 isnecessary. In this embodiment, the same electrical signal provided tothe RF modem modules 1052 is applied to the corresponding protectionmodule 1054, thus only a control signal indicating which module is to beused for transmission or reception is required for the one-to-oneprotection.

What is claimed is:
 1. A remote communication device for operation in abidirectional communication system, the device comprising: at least onesymbol mapper for mapping symbols using quadrature amplitude modulation(QAM) symbol mapping and phase-shift keying (PSK) symbol mapping; anInverse Fast Fourier Transform (IFFT) engine operating on frequencydomain that receives a frame of parallel frequency domain data based onsymbol data generated by the symbol mapper, the Inverse Fast FourierTransform (IFFT) engine generating a frame of time domain in-phase andquadrature phase data from the frame of parallel frequency domain datawhich is generated based on symbol data generated by the symbol mapper;at least one converter generating an analog signal based on the streamof time domain in-phase and quadrature phase data; a radio frequencytransmitter for transmitting a radio frequency signal based on theanalog signal; and a processor, wherein the symbol mapper is responsiveto the processor, and the processor controls synchronization of symboltiming and carrier frequency of transmissions from the radio frequencytransmitter.
 2. The remote communication device of claim 1, wherein theradio frequency transmitter modulates a carrier frequency with the atleast one analog signal.
 3. The remote communication device of claim 1,further comprising: a buffer receiving symbols of both payload andcontrol data from the at least one symbol mapper.
 4. The remotecommunication device of claim 1, wherein the at least one convertercomprises at least one digital to analog converter (DAC) receiving thetime domain in-phase and quadrature phase data and generating at leastone analog signal based on the stream of time domain in-phase andquadrature phase data.
 5. The remote communication device of claim 1,wherein the at least one symbol mapper comprises a binary phase-shiftkeying (BPSK) symbol mapper.
 6. A remote communication device for abi-directional communication system, the device comprising: a modem thatincludes: a symbol mapper that produces at least one stream of symbols;an Inverse Fast Fourier Transform (IFFT) engine operating on frequencydomain that receives a frame of parallel frequency domain data which isgenerated based on symbol data generated by the symbol mapper, theInverse Fast Fourier Transform (IFFT) engine generating a frame of timedomain in-phase and quadrature phase samples from the symbol datagenerated by the symbol mapper; at least one converter receiving thetime domain in-phase and quadrature phase samples and generating ananalog signal based on the time domain samples; and an amplifier thatamplifies for transmission a radio frequency carrier carrying the atleast one analog signal; and a processor that synchronizes thetransmissions from the amplifier for combining with other signals. 7.The remote communication device of claim 6, wherein the processoradjusts symbol timing.
 8. The remote communication device of claim 7,wherein the processor adjusts transmission characteristics align symbolsreceived at a host unit within a guard interval.
 9. A remotecommunication device, the device comprising: a symbol mapper thatproduces at least one stream of symbols; an Inverse Fast FourierTransform (IFFT) engine operating on frequency domain that receives aframe of parallel frequency domain data which is generated based onsymbol data generated by the symbol mapper, the Fast Fourier Transform(FFT) engine generating a frame of time domain in-phase and quadraturephase samples from the symbol data generated by the symbol mapper; atleast one converter receiving the time domain in-phase and quadraturephase samples and generating an analog signal based on the time domainsamples; and an amplifier that amplifies for transmission a radiofrequency carrier carrying the at least one analog signal; a processorthat causes transmissions from the amplifier to be orthogonal whenreceived at a multipoint-to-point host unit.
 10. The remotecommunication device of claim 9, further comprising a process betweenthe symbol mapper and the inverse Fast Fourier Transform engine, theprocess receiving symbol data produced by the symbol mapper andgenerating the frame of parallel frequency domain data.
 11. The remotecommunication device of claim 9, wherein the IFFT engine performs aninverse Fast Fourier Transform on the frame of parallel frequency domaindata to produce the time domain samples.
 12. The remote communicationdevice of claim 9, wherein the processor causes the transmissions to beorthogonal with respect to other signals when received at themultipoint-to-point host unit.